Semiconductor integrated circuit device and power supply system

ABSTRACT

A semiconductor integrated circuit device includes a first voltage terminal, a second voltage terminal, an output terminal, a high-side MOSFET connected between the first voltage terminal and the output terminal, a low-side MOSFET connected between the output terminal and the second voltage terminal and having first and second gate electrodes, a drive circuit that complementally switches on and off the high-side MOSFET and low-side MOSFET, and a second gate electrode control circuit that generates a second gate control signal supplied to the second gate electrode of the low-side MOSFET. The second gate electrode control circuit has a voltage generating circuit that supplies a negative voltage negative in polarity relative to a voltage at the source of the low-side MOSFET, to the second gate electrode of the low-side MOSFET.

TECHNICAL FIELD

The present invention relates to a semiconductor integrated circuitdevice and a power supply system, and relates to, for example, asemiconductor integrated circuit device used for voltage conversion anda power supply system equipped with the semiconductor integrated circuitdevice.

BACKGROUND

A power supply system that coverts a DC voltage having a given voltagevalue into a DC voltage having a voltage value different from the givenvoltage value is known as so-called DC/DC converter. A DC/DC converteris used in various electronic devices, one example of which is acomputer. In a computer, for example, a supply voltage supplied to amicroprocessor (hereinafter “CPU”) is generated by a DC/DC converter.

A power supply system like a DC/DC converter has multiple switchelements that cyclically change the direction of a current flowing fromone coil to another coil. Each of the multiple switch elements iscomposed of a transistor, such as metal-oxide-semiconductor field effecttransistor (hereinafter “MOSFET”). Various structures of the MOSFET usedin the DC/DC converter have been proposed. Patent Documents 1 and 2disclose various examples of MOSFET structures. In Patent Document 1, aMOSFET structure is shown in, for example, FIG. 1D in which a first gateelectrode 10 and a second gate electrode 12 are stacked vertically. InPatent Document 2, another MOSFET structure is shown in, for example,FIG. 4K in which a gate electrode 26 (hereinafter “first gateelectrode”) and a gate electrode 30 (hereinafter “second gateelectrode”) are stacked vertically.

PRIOR ART DOCUMENTS Patent Documents

Patent Document 1: Japanese Patent Application Laid-Open Publication No.63-296282

Patent Document 2: International Publication No. WO00/25365

SUMMARY

As shown in Patent Documents 1 and 2, the first gate electrode and thesecond gate electrode are stacked vertically. This structure forexample, as stated in Patent Document 1, allows reducing a capacitancebetween the second gate electrode and the drain region of the MOSFETwhile maintaining the highly integrated configuration, thereby improvingthe high-frequency characteristics of the MOSFET. Improving thehigh-frequency characteristics reduces the loss of the DC/DC converter,thus improving its efficiency. Improving the efficiency of the DC/DCconverter leads to a reduction in power consumption by an electronicdevice using the DC/DC converter, and is therefore important matter toachieve.

Before application of the invention, the inventor studied about such aDC/DC converter using the MOSFET having the first and second gateelectrodes. During the course of study, the inventor examined PatentDocuments 1 and 2.

According to Patent Document 1, the first gate electrode of the MOSFETis set to a given positive voltage and an input signal is supplied tothe second gate electrode of the same. According to Patent Document 2,the first gate electrode is connected to the source of the MOSFET. Aftera series of examinations, the inventor found that changing a voltagesupplied to the first gate electrode to a positive voltage or negativevoltage relative to a voltage supplied to the source of the MOSFETresults in a change in the characteristics (on-resistance value,capacitance value) of the MOSFET. The present invention has been madebased on this discovered knowledge.

The object of the present invention is to provide a power supply systemthat allows an improvement in overall efficiency and a semiconductorintegrated circuit device used in the power supply system.

The above and other preferred aims and novel characteristics of thepresent invention will be apparent from the description of the presentspecification and the accompanying drawings.

This specification of the present application discloses multiple meansfor solving problems. A typical means for solving problems will bedescribed from the viewpoint of a semiconductor integrated circuitdevice used in a power supply system and of a typical power supplysystem. In the following description, cases of using an n-channel typeMOSFET as a MOSFET will be explained. Obviously, a p-channel type MOSFETmay also be used as a MOSFET, in which case a potential relation of thevoltage is modified.

<Viewpoint of Semiconductor Integrated Circuit Device>

(1) A semiconductor integrated circuit device includes a first voltageterminal, a second voltage terminal, an output terminal, a first MOSFETconnected between the first voltage terminal and the output terminal,and a second MOSFET connected between the second voltage terminal andthe output terminal. The first MOSFET has a first input electrode, adrain, and a source, while the second MOSFET has a first inputelectrode, a drain, a source, and a second input electrode disposedcloser to the drain than the first input electrode.

An input signal that complementally switches on and off the first MOSFETand the second MOSFET is supplied to respective first input electrodesof the first MOSFET and second MOSFET. As a result, a current issupplied from the first voltage terminal and from the second voltageterminal alternately to the output terminal.

Meanwhile, a negative voltage negative in polarity relative to a voltageat the source of the second MOSFET is supplied to the second inputelectrode disposed closer to the drain than the first input electrode inthe second MOSFET. This further reduces a parasitic capacitance createdbetween the first input electrode and the drain. A reduction in theparasitic capacitance shortens an on-off shift time that the secondMOSFET takes when shifting from its off-state to on-state or vice versa.The first MOSFET and second MOSFET are switched on and offcomplementary, and during an on-off shift time required for the MOSFETsto shift from its off-state to on-state or vice versa, a current flowsthrough a path between the first voltage terminal and the second voltageterminal or between the first voltage terminal and the output terminalor between the second voltage terminal and the output terminal. Byreducing this on-off shift time, the loss (power consumption) of thesemiconductor integrated circuit device is reduced.

(2) According to one embodiment, the second input electrode and thefirst input electrode of the second MOSFET are formed of a second metallayer and a first metal layer embedded in a first semiconductor regionand a third semiconductor region stacked together, respectively. Thefirst semiconductor region makes up the drain of the second MOSFET,while the third semiconductor region makes up the source of the secondMOSFET. Hence, the second input electrode is disposed closer to thedrain of the second MOSFET than the first input electrode. Being stackedtogether, the electrodes are packaged into a highly integratedstructure. In this case, the channel of the second MOSFET is created ina second semiconductor region interposed between the first semiconductorregion and the third semiconductor region.

(3) According to one embodiment, the semiconductor integrated circuitdevice has a selecting circuit that selects a positive voltage ornegative voltage positive or negative in polarity relative to a voltageat the source of the second MOSFET. By supplying a positive voltagepositive in polarity relative to a voltage at the source to the secondinput electrode of the second MOSFET, the on-resistance of the secondMOSFET is reduced when the MOSFET is switched on. Reducing theon-resistance of the second MOSFET enables a reduction in the loss ofthe second MOSFET.

Selecting the polarity of a voltage supplied to the second inputelectrode by the selecting circuit (with respect to a voltage at thesource as a reference voltage) allows selection between loss reductionby shortening the on-off shift time and loss reduction by reducing theon-resistance.

(4) According to one embodiment, the selecting circuit selects thepolarity of a voltage supplied to the second input electrode, insynchronization with switching on and off of the second MOSFET by aninput signal supplied to the first input electrode of the second MOSFET.As a result, loss reduction by shortening the on-off shift time and lossreduction by reducing the on-resistance can be performed insynchronization with switching on and off of the second MOSFET.

(5) According to one embodiment, the semiconductor integrated circuitdevice has a detecting circuit that detects a current flowing throughthe output terminal to determine whether the value of the currentexceeds a given current value. In response to a detection signal fromthe detecting circuit, the selecting circuit changes the polarity of avoltage supplied to the second input electrode. Hence, either lossreduction by shortening the on-off shift time or loss reduction byreducing the on-resistance can be selected according to the value of aload current required by a load connected to the output terminal. Inother words, a voltage polarity for proper loss reduction is selectedaccording to the size of the load current. This allows loss reductionaccording to the size of the load current.

The load current varies depending on the condition of the load. Throughthe above configuration, therefore, a proper loss reduction means(shortening the on-off shift time or reducing the on-resistance) can beselected according to the load condition.

(6) According to one embodiment, a semiconductor integrated circuitdevice includes a first voltage terminal, a second voltage terminal, anoutput terminal, a first MOSFET connected between the first voltageterminal and the output terminal, and a second MOSFET connected betweenthe second voltage terminal and the output terminal. Each of the firstMOSFET and second MOSFET has a first input electrode, a drain, a source,and a second input electrode disposed closer to the drain than the firstinput electrode.

The first MOSFET and the second MOSFET are complementally switched onand off by an input signal. As a result of complemental switching on andoff of the MOSFETs, a current is supplied from the first voltageterminal and from the second voltage terminal alternately to the outputterminal. The value of a current flowing through the output terminalchanges depending on a load current required by a load connected to theoutput terminal.

The semiconductor integrated circuit device further includes a detectingcircuit that detects the value of a current flowing through the outputterminal and a control circuit that in response to a detection signalfrom the detecting circuit, supplies different voltages to respectivesecond input electrodes of the first MOSFET and the second MOSFET.

As a result, according to the value of the load current, loss reductionby shortening an on-off time and loss reduction or loss increasesuppression by reducing on-resistance or suppressing an on-resistanceincrease are carried out at the first MOSFET and the second MOSFET. Thisallows a reduction in the loss of the semiconductor integrated circuitdevice according to the load condition, thereby allowing an improvementin efficiency.

According to one embodiment, when the value of a current flowing throughthe output terminal exceeds a given current value, the control circuitsupplies a positive voltage positive in polarity relative to a voltageat respective sources of the first MOSFET and the second MOSFET, torespective second gate electrode of the first MOSFET and the secondMOSFET. When the value of the current flowing through the outputterminal does not exceed the given current value, the control circuitsupplies a negative voltage negative in polarity relative to a voltageat respective sources of the first MOSFET and the second MOSFET, torespective second gate electrode of the first MOSFET and the secondMOSFET. Hence, loss reduction by shortening the on-off shift time andloss reduction by reducing the on-resistance are carried out accordingto the value of the output current (load current) flowing through theoutput terminal.

<Viewpoint of Typical Power System>

In multiple embodiments of the power supply system, the power supplysystem includes the semiconductor integrated circuit device and a coilelement. One end of the coil element is connected to the output terminalof the semiconductor integrated circuit device, and a current suppliedfrom the output terminal to the coil element changes in directioncyclically.

In each of the embodiments of the power supply system, the semiconductorintegrated circuit device has any one of the means for solving theproblems (1) to (6) described in <Viewpoint of Semiconductor IntegratedCircuit Device>. Each of the means for solving the problems (1) to (6)allows the semiconductor integrated circuit device to reduce its loss,thus allowing the power supply system to reduce its loss, therebyimproving overall efficiency.

In the power supply system equipped with the semiconductor integratedcircuit device having the means for solving the problems (5) or (6), adevice (e.g., CPU) supplied with power from the power supply system isregarded as a load. A current required by the load (load current)changes depending on the operation status of the device regarded as theload.

Examinations conducted by the inventor of the present application hasrevealed that with regard to the total loss of the semiconductorintegrated circuit device, the ratio between a loss caused by the on-offshift time and a loss caused by the on-resistance in a case where theload is heavy (heavy load) and the load current is high is differentfrom the ratio between the same in a case where the load is light (lightload) and the load current is relatively low. The inventor's examinationhas lead to a conclusion that the ratio of the loss caused by theon-resistance becomes higher as the load current becomes higher.

According to the means for solving the problems (5) or (6), the loadcurrent is detected by the detecting circuit, and in response to adetection signal, the selecting circuit (means (5)) or the controlcircuit (means (6)) (the second MOSFET according to the means (5) andthe first and second MOSFETs according to the means (6)) selects avoltage supplied to the second input electrode of the MOSFET. When thevalue of the load current exceeds the given current value, the selectingcircuit or control circuit selects a positive voltage positive inpolarity relative to a voltage at the source of the MOSFET, as thevoltage is supplied to the second input electrode of the MOSFET. Thisreduces loss that arises when the load is heavy. When the load is light,in contrast, the selecting circuit or control circuit selects a negativevoltage negative in polarity relative to a voltage at the source of theMOSFET, and supplies the selected voltage to the second input electrodeof the MOSFET. This reduces loss that arises when the load is light.

In this manner, according to the condition of a device serving as theload, loss is reduced by a selected method effective for loss reduction(loss reduction by shortening the on-off shift time or loss reduction byreducing the on-resistance). As a result, the loss of the power supplysystem can be reduced according to the load condition.

It is understood from the above description that the first inputelectrode corresponds to, for example, the second gate electrode ofPatent Document 1 and the second input electrode corresponds to thefirst gate electrode of the same. In the following description, losscaused during the on-off shift time is also referred to as switchingloss and loss caused by the on-resistance is also referred to asconduction loss.

According to one embodiment, a power supply system that can improveoverall efficiency and a semiconductor integrated circuit device used inthe power supply system are provided.

BRIEF DESCRIPTIONS OF THE DRAWINGS

FIGS. 1(A) to 1(D) are circuit diagrams and voltage waveform chartsshowing principle circuits of a semiconductor integrated circuit deviceaccording to a first embodiment;

FIGS. 2(A) and 2(B) are block diagrams and circuit diagrams showingconfigurations of principle parts of a semiconductor integrated circuitdevice according to a second embodiment;

FIGS. 3(A) to 3(D) are waveform charts showing an operation of thesemiconductor integrated circuit device according to the secondembodiment;

FIGS. 4(A) to 4(D) are explanatory diagrams of an operation of thesemiconductor integrated circuit device according to the secondembodiment;

FIG. 5 is a block diagram showing a configuration of a principle part ofa semiconductor integrated circuit device according to a thirdembodiment;

FIGS. 6(A) and 6(B) are waveform charts showing an operation of thesemiconductor integrated circuit device according to the thirdembodiment;

FIG. 7 is a block diagram showing a configuration of a principle part ofa semiconductor integrated circuit device according to a fourthembodiment;

FIGS. 8(A) to 8(D) are waveform charts showing an operation of thesemiconductor integrated circuit device according to the fourthembodiment;

FIGS. 9(A) to 9(F) are explanatory diagrams of an operation of thesemiconductor integrated circuit device according to the fourthembodiment;

FIG. 10 is a characteristic diagram showing the characteristics of thesemiconductor integrated circuit device according to the fourthembodiment;

FIGS. 11(A) to 11(F) are explanatory diagrams of an operation of thesemiconductor integrated circuit device according to the firstembodiment;

FIGS. 12(A) to 12(F) are explanatory diagrams of an operation of thesemiconductor integrated circuit device according to the firstembodiment;

FIG. 13 is a characteristic diagram showing the characteristics of thesemiconductor integrated circuit device according to the firstembodiment;

FIGS. 14(A) to 14(F) are explanatory diagrams of an operation of thesemiconductor integrated circuit device according to the secondembodiment;

FIG. 15 is a characteristic diagram showing the characteristics of thesemiconductor integrated circuit device according to the secondembodiment;

FIGS. 16(A) and 16(B) are block diagrams and waveform charts showingconfigurations of principle parts of a semiconductor integrated circuitdevice according to a fifth embodiment;

FIGS. 17(A) to 17(E) are explanatory diagrams of an operation of thesemiconductor integrated circuit device according to the fifthembodiment;

FIG. 18 is a characteristic diagram showing the characteristics of thesemiconductor integrated circuit device according to the fifthembodiment;

FIGS. 19(A) and 19(B) are diagrams and waveform charts showingconfigurations of principle parts of a semiconductor integrated circuitdevice according to a sixth embodiment;

FIGS. 20(A) to 20(E) are explanatory diagrams of an operation of thesemiconductor integrated circuit device according to the sixthembodiment;

FIG. 21 is a characteristic diagram showing the characteristics of thesemiconductor integrated circuit device according to the sixthembodiment;

FIGS. 22(A) and 22(B) are block diagrams and waveform charts showingconfigurations of principle parts of a semiconductor integrated circuitdevice according to a seventh embodiment;

FIGS. 23(A) to 23(E) are explanatory diagrams of an operation of thesemiconductor integrated circuit device according to the seventhembodiment;

FIG. 24 is a characteristic diagram showing the characteristics of thesemiconductor integrated circuit device according to the seventhembodiment;

FIG. 25 is a waveform chart showing the waveforms of signals from aprinciple part of a semiconductor integrated circuit device according toan eighth embodiment;

FIG. 26 is a characteristic diagram showing the characteristics of thesemiconductor integrated circuit device according to the eighthembodiment;

FIG. 27 is a block diagram showing a configuration of a semiconductorintegrated circuit device according to a ninth embodiment;

FIG. 28 is a block diagram showing a second modification example of thesemiconductor integrated circuit device according to the ninthembodiment;

FIG. 29 is a block diagram showing a third modification example of thesemiconductor integrated circuit device according to the ninthembodiment;

FIG. 30 is a block diagram showing a configuration of a semiconductorintegrated circuit device according to a tenth embodiment;

FIG. 31 is a circuit diagram showing a configuration of a principle partof the semiconductor integrated circuit device according to the tenthembodiment;

FIGS. 32(A) to 32(E) are waveform charts showing an operation of thesemiconductor integrated circuit device according to the tenthembodiment;

FIG. 33 is a block diagram showing a configuration of a semiconductorintegrated circuit device according to an eleventh embodiment;

FIG. 34 is a circuit diagram showing a configuration of a principle partof the semiconductor integrated circuit device according to the eleventhembodiment;

FIGS. 35(A) to 35(E) are waveform charts showing an operation of thesemiconductor integrated circuit device according to the eleventhembodiment;

FIGS. 36(A) and 36(B) are diagrammatical views and plan views showingthe relation between the semiconductor integrated circuit device, apackage, and the power supply system;

FIGS. 37(A) and 37(B) are plan views and sectional views of a MOSFEThaving a first gate electrode and a second gate electrode;

FIG. 38 is an explanatory diagram of the loss of the semiconductorintegrated circuit device;

FIGS. 39(A) and 39(B) are explanatory diagrams of the loss of theMOSFET;

FIG. 40 is a block diagram showing configurations of the semiconductorintegrated circuit device and the power supply system according to theembodiments; and

FIGS. 41(A) to 41(G) are waveform charts showing an operation of thesemiconductor integrated circuit device according to the embodiments.

DESCRIPTIONS OF THE PREFERRED EMBODIMENTS

Hereinafter, embodiments of the present invention will be described indetail with reference to the accompanying drawings. Note that componentshaving the same function are denoted by the same reference symbolsthroughout the drawings for describing the embodiment, and therepetitive description thereof will be omitted.

<Outline of Power Supply System and Semiconductor Integrated CircuitDevice Used in Power Supply System>

Multiple embodiments will hereinafter be described in order. An outlineof configurations and operations of a power supply system and asemiconductor integrated circuit device used in the power supply system,the power supply system and semiconductor integrated circuit deviceapplying in common to the embodiments, will first be described.

FIG. 40 is a block diagram showing a configuration of a power supplysystem. In FIG. 40, 4000 denotes the power supply system and 4001denotes a load connected to the power supply system 4000. The load 4001can be regarded as an equivalent to a current source and is thereforedenoted by a current source symbol. As describe above, for example, aCPU is equivalent to the load. The power supply system 4000 includes acontrol semiconductor integrated circuit device 4007, a coil element4008, a smoothing capacitor 4009, a boot capacitor 4010, and multiplesemiconductor chips 4003 to 4006 sealed in a single package 4002.Components making up the power supply system 4000 are not limited tothese components.

In this embodiment, the semiconductor chips sealed in the single package4002 are a high-side MOSFET 4005, a low-side MOSFET 4006, and a driver4003 that drives the high-side MOSFET 4005 and the low-side MOSFET 4006.In other words, each of the high-side MOSFET 4005, the low-side MOSFET4006, and the driver 4003 is formed as a separate semiconductor chip.These three semiconductor chips are sealed in the single package, whichwill be described later referring to FIGS. 36(A) and 36(B). Since a unitof the package is mounted on, for example, a printed board, etc., thepackage 4002 is referred to as semiconductor integrated circuit devicein this specification of the present application. In the followingdescription, therefore, the package 4002 is explained as thesemiconductor integrated circuit device. According to this embodiment, avoltage supplied to the drain of the MOSFET 4005 is higher than avoltage supplied to the source of the MOSFET 4006. For this reason, theMOSFET 4005 is referred to as high-side MOSFET while the MOSFET 4006 isreferred to as low-side MOSFET.

In FIG. 40, each of T1 to T6 denotes a terminal formed on thesemiconductor integrated circuit device 4002. The semiconductorintegrated circuit device 4002 has multiple terminals out of whichprinciple terminals are shown as the terminals T1 to T6 in FIG. 40. Forexample, a terminal that receives an input signal from the controlsemiconductor integrated circuit device 4007 is omitted from FIG. 40.The terminal T1 is an output terminal that outputs an output signal VSWHfrom the semiconductor integrated circuit device 4002, the terminal T2is a voltage terminal that supplies a ground voltage PGND to thelow-side MOSFET 4006, and the terminal T3 is a voltage terminal thatsupplies a ground voltage CGND to the driver. The terminal T4 is avoltage terminal that supplies a supply voltage VCIN to a controlcircuit 4004, the terminal T5 is a voltage terminal that supplies asupply voltage BOOT corresponding to the voltage of the output signalVSWH, to the driver, and the terminal T6 is a voltage terminal thatsupplies an input voltage VIN to the high-side MOSFET 4005.

As described above, each of the high-side MOSFET 4005 and low-sideMOSFET 4006 is an n-channel type MOSFET, and has a first gate electrodeG1 equivalent to a first input electrode, a second gate electrode G2equivalent to a second input electrode, a source S, and a drain D, whichwill be described later referring to FIGS. 37(A) and 37(B). The secondinput electrode (second gate electrode) G2 is located closer to thedrain D than the first input electrode (first gate electrode) G1, andthe MOSFET is switched on and off according to a voltage supplied to thefirst gate electrode G1. Functioning as the n-channel type MOSFET, theMOSFET is switched on when a positive voltage positive in polarityrelative to a voltage at the source S and higher than a given voltage(threshold voltage) is supplied to the first gate electrode G1, while itis switched off when a voltage equal to or lower than the thresholdvoltage relative to the voltage at the source S is supplied to the firstgate electrode G1.

A positive voltage or negative voltage positive or negative in polarityrelative to a voltage at the source S as a reference voltage is suppliedto respective second input electrodes (second gate electrodes) G2 of thehigh-side MOSFET 4005 and low-side MOSFET 4006, which will be describedlater in multiple embodiments. Since the outline of the power supplysystem and the semiconductor integrated circuit device is describedhere, supply of the positive voltage or negative voltage to the secondinput electrodes will not be described further. At the high-side MOSFET4005 and the low-side MOSFET 4006 depicted in FIG. 40, a parasitic diodeis created between a semiconductor region in which the MOSFET is formedand a semiconductor region in which the drain is formed, and thisparasitic diode is denoted as DD. The back gates of these MOSFETs areconnected to the sources, respectively. Connection destinations forrespective second input electrodes G2 of the high-side MOSFET 4005 andlow-side MOSFET 4006 will be described later in each embodiment, and aretherefore not indicated in FIG. 40.

The high-side MOSFET 4005 has a source-drain path connected in seriesbetween the voltage terminal T6 and the output terminal T1 and has thefirst gate electrode G1 connected to an output terminal T7 of the driver4003. The low-side MOSFET 4006 has a source-drain path connected inseries between the output terminal T1 and the voltage terminal T2 andhas the first gate electrode G1 connected to an output terminal T9 ofthe driver 4003. According to the embodiment, the ground voltage PGND issupplied to the voltage terminal T2, and a positive voltage higher thanthe ground voltage PGND is supplied as the input voltage VIN, to thevoltage terminal T6. In FIG. 40, therefore, the drain D of the high-sideMOSFET 4005 is connected to the voltage terminal T6 and the source S ofthe same is connected to the output terminal T1. Likewise, the drain Dof the low-side MOSFET 4006 is connected to the output terminal T1 andthe source S of the same is connected to the voltage terminal T2.

An output signal (drive signal) GH output from the output terminal T7 ofthe driver 4003 is input to the first gate electrode G1 of the high-sideMOSFET 4005 that receives the incoming signal GH as an input signal. Anoutput signal (drive signal) GL output from the output terminal T9 ofthe driver 4003 is input to the first gate electrode G1 of the low-sideMOSFET 4006 that receives the incoming signal GL as an input signal. Thedriver 4003 changes the voltages of the drive signals GH and GL so thatthe high-side MOSFET 4005 and the low-side MOSFET 4006 are switched onand off complementally. As a result of complemental switching on and offof the high-side MOSFET 4005 and the low-side MOSFET 4006 caused by thedrive signals GH and GL, the voltage terminal VIN or the voltageterminal PGND is connected electrically to the output terminal T1through the source-drain path of the high-side MOSFET 4005 or thelow-side MOSFET 4006.

When the high-side MOSFET 4005 is switched on, a current is suppliedfrom the input voltage VIN node through the output terminal T1 to oneend of the coil element 4008. When the low-side MOSFET 4006 is switchedon, on the contrary, a current is supplied from the one end of the coilelement 4008 toward the output terminal T1 and is finally supplied tothe voltage terminal T2. Repeated complemental switching on and off ofthe high-side MOSFET 4005 and the low-side MOSFET 4006 results in supplyof a cyclically alternating current to the one end of the coil element4008, thus generating a counter electromotive force, which creates anoutput voltage Vout on the other end of the coil element 4008, theoutput voltage Vout being different in voltage value from the inputvoltage VIN. This output voltage Vout is smoothed out by the smoothingcapacitor 4009 and is supplied to the load 4001.

Meanwhile, a voltage generated on the one end of the coil element 4008is supplied to a boot capacitor 4010. As a result of cyclic changes ofthe voltage at the one end of the coil element 4008, the boot capacitor4010 generates a voltage BOOT higher than a voltage at the outputterminal T1 and supplies the voltage BOOT to the voltage terminal T5.

The driver 4003 has drive circuits 4011 and 4012 and the control circuit4004. The drive circuit 4011 operates with a voltage from a voltageterminal T8 as a reference voltage and a voltage BOOT from the voltageterminal T5 as a supply voltage, and outputs a signal following anoutput signal f from the control circuit 4004, as the drive signal GH.Because the voltage terminal T8 is connected to the source S of thehigh-side MOSFET 4005, the drive circuit 4011 outputs the drive signalGH to which a voltage at the source S of the high-side MOSFET 4005, thatis, a voltage at the output terminal T1 serves as a reference voltage.Hence, the voltage of the drive signal GH changes between, for example,the reference voltage (voltage at the voltage terminal T8) and thevoltage BOOT.

The drive circuit 4012 operates with a voltage at a voltage terminal T10as a reference voltage and the supply voltage VCIN supplied to thevoltage terminal T4 as a supply voltage. Because the voltage terminalT10 is connected to the voltage terminal 12, the drive circuit 4012operates with the ground voltage PGND and the voltage VCIN as supplyvoltages and outputs the drive signal GL following an output signal cfrom the control circuit 4004, to the first gate electrode G1 of thelow-side MOSFET 4006 via the output terminal T9.

The control circuit 4004 operates with the ground voltage CGND suppliedto the voltage terminal T3, the voltage VCIN supplied to the voltageterminal T4, and the voltage BOOT supplied to the voltage terminal T5that serve as operation power supply. The control circuit 4004 hascontrol terminals T11 and T12, and generates the output signals f and cfollowing a pulse width control signal PWM (input signal a) suppliedfrom the control semiconductor integrated circuit device 4007 to thecontrol terminal T11. To the control terminal T12, a control signalDISBL# that gives an instruction on whether or not to operate thecontrol circuit 4004 is supplied. In FIG. 40, the supply voltage VCIN issupplied to the control terminal T12 via a resistance element 4022.Because the supply voltage VCIN is a high voltage, the control signalDISBL# is a high-voltage signal, in which case the control circuit 4004generates output signals g and c following the pulse width controlsignal PWM supplied to the control terminal T11. When the control signalDISBL# is shifted to a low-voltage signal, the control circuit 4004 isshifted to a non-operation state. In this manner, the power supplysystem 4000 can be controlled to its operation state or non-operationstate through the control signal DISBL#.

The ground voltage CGND supplied to the voltage terminal T3 of thecontrol circuit 4004 is substantially the same voltage (ground voltageGND) as the ground voltage PGND supplied to the voltage terminal T2connected to the source S of the low-side MOSFET 4006. In thisembodiment, the voltage terminal T3 that supplies the ground voltage tothe control circuit 4004 is electrically isolated from the voltageterminal T2 that supplies the ground voltage to the source S of thelow-side MOSFET 4006. This prevents, for example, a change in the groundvoltage PGND caused by the operation of the low-side MOSFET 4006 frombeing transmitted to the control circuit 4004. The value of the voltageBOOT supplied to the voltage terminal T5 is determined to be higher thanthe value of the source voltage VCIN supplied to the voltage terminalT4. The control circuit 4004 is so configured that the high-levelvoltage of the output signal f from the control circuit 4004 matches thevoltage BOOT. Hence, the output signal f having the voltage BOOT higherthan the source voltage VCIN is input to the drive circuit 4011.

The input voltage VIN is, for example, 12 V and the source voltage VCINfor the driver 4003 is, for example, 5 V. The voltage values of theinput voltage VIN and the source voltage VCIN are not limited to thesevalues.

In this embodiment, each of the drive circuits 4011 and 4012 functionsas a buffer circuit. The drive circuit 4012 thus supplies the outputsignal c from the control circuit 4004 to the first gate electrode G1 ofthe low-side MOSFET 4006. The voltage of the drive signal GL supplied tothe first gate electrode G1 of the low-side MOSFET 4006 shifts betweenthe source voltage VCIN and the ground voltage.

The drive circuit 4011 supplies the output signal f from the controlcircuit 4004 to the first gate electrode G1 of the high-side MOSFET4005. The drive circuit 4011 is supplied with the voltage BOOT as asupply voltage, so that the high-level voltage of the output signal fmatches the voltage BOOT. As a result, the voltage of the drive signalGH supplied to the first gate electrode G1 of the high-side MOSFET 4005shifts between the voltage BOOT and the voltage VSWH at the voltageterminal T8 (voltage at the source of the high-side MOSFET). In thismanner, by increasing the voltage of the drive signal GH supplied to thefirst gate electrode G1 of the high-side MOSFET 4005, the loss of thehigh-side MOSFET 4005 caused by the threshold voltage is reduced.

The output voltage Vout output from the output terminal T13 of the powersupply system 4000 is supplied to the control semiconductor integratedcircuit device 4007. The control semiconductor integrated circuit device4007 generates the pulse width control signal PWM (input signal a)according to the value of the output voltage Vout, that is, generatesthe pulse width control signal PWM having a pulse width (e.g., periodduring which the signal's voltage level stays high) corresponding to thevalue of the output voltage Vout.

The control circuit 4004 in this embodiment has level shifters 4013 and4014, an input logical circuit 4018, AND circuits 4016 and 4017, a NORcircuit 4014, an inverter circuit 4020, a low-voltage detecting circuit4021, and an overlap preventing circuit 4015. The level shifter 4013 issupplied with the voltage BOOT so that the high-level voltage of theoutput signal f output from the level shifter 4013 matches the voltageBOOT. The overlap preventing circuit 4015 is provided with a voltageconverting circuit that converts the high-level voltage of the outputsignal f to a proper voltage. Providing the voltage converting circuit,however, is not absolute necessity.

It is indicated in FIG. 40 that the AND circuit 4016 is a three-inputAND circuit having one reverse input terminal and two non-reverse inputterminals and that the AND circuit 4017 is a three-input AND circuithaving two reverse input terminals and one non-reverse input terminal.The reverse input terminal is the terminal that reverses an incomingsignal in polarity and that supplies the reversed signal to the ANDcircuit. The non-reverse terminal is the terminal that supplies anincoming signal left as it is to the AND circuit. In this embodiment,the inverter 4020 has a hysteresis function for preventing noise-causedmalfunctioning.

The input logical circuit 4018 has multiple functions, which will not bedescribed. The pulse width control signal PWM (input signal a) outputfrom the control semiconductor integrated circuit device 4007 travelsthrough the control terminal T11 to enter the input logical circuit4018. An output signal b from the input logical circuit 4018 is suppliedto one reverse-input terminal of the AND circuit 4017 and to onenon-reverse input terminal of the AND circuit 4016. An output signalfrom the NOR circuit 4019 is supplied to the other non-reverse inputterminal of the AND circuit 4016 and to the non-reverse input terminalof the AND circuit 4017. One input terminal of the NOR circuit 4019 issupplied with the control signal DISBL# transmitted thereto via theinverter 4019, while the other input terminal of the same is suppliedwith the source voltage VCIN transmitted thereto via the low-voltagedetecting circuit 4021.

The source voltage VCIN is input to the low-voltage detecting circuit4021. When the source voltage VCIN drops below a given voltage, thelow-voltage detecting circuit 4021 generates a high-voltage level outputsignal and supplies it to the NOR circuit 4019. When the control signalDISBL# is shifted to a low-voltage level signal or the source voltageVCIN drops below the given voltage, therefore, the NOR circuit 4019generates a low-voltage level output signal. When the output signal fromthe NOR circuit 4019 goes low in voltage level, the AND circuits 4016and 4017 do not transmit the output signal b from the input logicalcircuit 4018 or/and the output signals d and g from the overlappreventing circuit 4015, to the output terminals of the AND circuits4016 and 4017, respectively. As a result, the control circuit 4004shifts to a non-operation state. In contrast, when the source voltageVCIN is higher than the given voltage and the high-voltage level controlsignal DISBL# is supplied, an output signal from the NOR circuit 4019goes high in voltage level. As a result, the AND circuits 4016 and 4017transmit the output signal b from the input logical circuit 4018 or/andthe output signals d and g from the overlap preventing circuit 4015, tothe output terminals of the AND circuits 4016 and 4017, respectively,which brings the control circuit 4004 into an operation state.

The reverse input terminal of the AND circuit 4016 is supplied with theoutput signal d from the overlap preventing circuit 4015, and the otherreverse input terminal of the AND circuit 4017 is supplied with theoutput signal g from the overlap preventing circuit 4015. An outputsignal e from the AND circuit 4016 is transmitted through the levelshifter 4013 and is input as the output signal f from the controlcircuit 4004, to the drive circuit 4011. This output signal f is inputalso to the overlap preventing circuit 4015. Meanwhile, the outputsignal c from the AND circuit 4017 is input as an output signal from thecontrol circuit 4004, to the drive circuit 4012. The output signal c isshifted in voltage level by the level shifter 4014 to a signal with adesired voltage and is supplied to the overlap preventing circuit 4015.

The overlap preventing circuit 4015 is the circuit that prevents thehigh-side MOSFET 4005 and low-side MOSFET 4006 from switching onsimultaneously. To prevent both MOSFETs from switching onsimultaneously, the overlap preventing circuit 4015 receives the outputsignals c and f and generates the output signals d and g whosehigh-voltage level periods do not overlap each other. Such a circuit canbe constructed by, for example, combining together multiple logicalcircuits and delay circuits.

The operation of the power supply system of FIG. 40 will then bedescribed, referring to FIGS. 41(A) to 41(G), which are waveform chartsof the above signals (output signals and control signals) a to g. InFIGS. 41(A) to 41(G), the horizontal axis represents time and thevertical axis represents voltage.

FIG. 41(A) is a waveform chart of the pulse width control signal PWM(denoted by ‘a’ in FIG. 41(A)) output from the control semiconductorintegrated circuit device 4007. In FIG. 41(A), for simpler explanation,only one period during which the signal remains high in voltage level(pulse width) is shown. Such periods during which the signal remainshigh in voltage level (pulse width), however, arise cyclically.According to this embodiment, the control semiconductor integratedcircuit device 4007 changes the pulse width (period during which thesignal remains high in voltage level) of the pulse width control signalPWM according to the value of the output voltage Vout from the powersupply system 4000 and regulates the value of the output voltage Vout toa given value. The control semiconductor integrated circuit device 4007performs such control by adopting so-called PWM control method.

In FIG. 41(A), the pulse width control signal PWM (a) at its low-voltagelevel shifts to a high-voltage level at time t1. In response to thischange of the pulse width control signal PWM, the input logical circuit4018 causes the output signal b at its low-voltage level to shift to ahigh-voltage level at time t2 after an elapse of a given delay time fromtime t1 (FIG. 41(B)). In response to the shift of the output signal b toits high-voltage level, the output signal b being supplied to thereverse input terminal of the AND circuit 4017, the AND circuit 4017causes the output signal c to shift to a low-voltage level at time t2(FIG. (C)). In response to the shift of the output signal c to itslow-voltage level, the drive circuit 4012 generates the low-voltagelevel drive signal GL and supplies it to the first gate electrode G1 ofthe low-side MOSFET 4006. Because the drive circuit 4012 functions asthe buffer, the drive signal GL and the output signal c synchronize witheach other. Therefore, the waveform depicted in FIG. 41(C) can beregarded as the waveform of the drive signal GL. Hence reference numeralGL representing the drive signal GL is indicated as “(GL)” in FIG.41(C).

The overlap preventing circuit 4015 receives the output signal c havingchanged in voltage level (from high-voltage level to low-voltage level)via the level shifter 4014. Upon receiving this output signal c havingchanged in voltage level, the overlap preventing circuit 4015 causes theoutput signal d at its high-voltage level to shift to a low-voltagelevel at time t3 after an elapse of a given delay time from time t2(FIG. 41(D)). Because the output signal d is supplied to the reverseinput terminal of the AND circuit 4016, the AND circuit 4016 causes theoutput signal e at its low-voltage level to shift to a high-voltagelevel at time t3, in response to the shift of the output signal d to itslow-voltage level (FIG. 41(E)).

When the output signal e from the AND circuit 4016 changes in voltagelevel from the low-voltage level to high-voltage level, the levelshifter 4013 generates the high-voltage level output signal f inresponse to the change in the voltage level of the output signal e.Because the level shifter 4013 is supplied with the voltage BOOT as anoperating voltage, the high-voltage level of the output signal f matchesthe voltage BOOT. This output signal f is transmitted through the drivecircuit 4011 functioning as the buffer to the first gate electrode G1 ofthe high-side MOSFET 4005. The drive circuit 4011 operates with thevoltage BOOT and the voltage VSWH at the voltage terminal T8. Because ofthis, the high-voltage level of the drive signal GH supplied from thedrive circuit 4011 to the first gate electrode G1 of the high-sideMOSFET 4005 matches the voltage BOOT, while the low-voltage level of thedrive signal GH matches the voltage VSWH. Because the output signal fand the drive signal GH synchronize with each other, FIG. 41(F) depictsthe output signal f and drive signal GH that go high in voltage level(voltage BOOT) at time t4.

When the output signal f shifts to a high-voltage level signal, theoverlap preventing circuit 4015 causes the output signal g at itslow-voltage level to shift to a high-voltage level at time t5 after anelapse of a given time from time t4 (FIG. 41(G)).

As described above, in response to the shift of the pulse width controlsignal PWM (denoted by a in FIG. 41) from its low-voltage level tohigh-voltage level, the drive signal GL from the drive circuit 4012shifts from its high-voltage level to low-voltage level at time t2. Attime t4 later than time t2, the drive signal GH from the drive circuit4011 shifts from its low-voltage level (VSWH) to high-voltage level(BOOT). Hence, the low-side MOSFET 4006 starts shifting to its off-stateat time t2, and the high-side MOSFET 4005 starts shifting to itson-state at time t4. As a result, the input voltage VIN is supplied tothe output terminal T1 via the high-side MOSFET 4005 and is thereforesupplied to the one end of the coil element 4008 and to one end of theboot capacitor 4010. In other words, a current is supplied from thevoltage terminal T6 to the one end of the coil element 4008 via theoutput terminal T1.

Subsequently, at time t6, the pulse width control signal PWM (denoted by‘a’ in FIG. 41(A)) at its high-voltage level shifts to a low-voltagelevel (FIG. 41(A)). In response to this voltage level change, the inputlogical circuit 4018 causes the output signal b at its high-voltagelevel to shift to a low-voltage level at time t7 (FIG. 41(B)). As aresult of the shift of the output signal b to the low-voltage level, theoutput signal e from the AND circuit 4016 shifts to a low-voltage levelat time t7 (FIG. 41(E)).

When the output signal e shifts to the low-voltage level, the levelshifter 4013 causes the output signal f at its high-voltage level(voltage BOOT) to shift to a low-voltage level. In response to thevoltage shift of the output signal f from the level shifter 4013, thedrive circuit 4011 causes the drive signal GH at its high-voltage level(BOOT) to shift to a low-voltage level (VSWH) (at time t8 in FIG.41(F)).

When the output signal f shifts from the high-voltage level to thelow-voltage level, the overlap preventing circuit 4015 causes the outputsignal g at its high-voltage level to shift to a low-voltage level attime t9 after an elapse of a given time from time t8, in response to theshift of the output signal f to the low-voltage level (FIG. 41(G)). As aresult, low-voltage level signals are supplied to the two reverse inputterminals of the AND circuit 4017, respectively, which causes the outputsignal c from the AND circuit 4017 to shifts from its low-voltage levelto high-voltage level (time t9 in FIG. 41(C)). This output signal c isheld temporarily by the drive circuit 4012 as a buffering action and issupplied to the first gate electrode G1 of the low-side MOSFET 4006 asthe drain signal GL.

When the output signal c shifts from the low-voltage level to thehigh-voltage level at time t9, the overlap preventing circuit 4015causes the output signal d at its low-voltage level to shift to ahigh-voltage level after an elapse of a given time from the time ofshift of the output signal c to the high-voltage level (time t10 in FIG.41(D)).

From time t10 onward, a waveform condition before time t1 results, andafter the pulse width control signal PWM goes high again in voltagelevel, the above operations are repeated.

As described above, the drive signal GH supplied to the first gateelectrode G1 of the high-side MOSFET 4005 shifts to the low-voltagelevel (VSWH) at time t8 and the drive signal GL supplied to the firstgate electrode G1 of the low-side MOSFET 4006 shifts to the low-voltagelevel at time t9 later than time t8. This means that the generation of aperiod during which the high-side MOSFET 4005 and low-side MOSFET 4006are switched on simultaneously can be prevented.

At time t9, the shift of the drive signal GL to the high-voltage level(VCIN) causes the low-side MOSFET 4006 to switch to its on-state. As aresult, the output terminal T1 is connected to the voltage terminal T2via the low-side MOSFET 4006. In other words, a current flows from theone end of the coil element 4008 toward the voltage terminal T2 via theoutput terminal T1.

Being supplied with an alternating current, the coil element 4008generates a counter electromotive force and also generates the outputvoltage Vout different in voltage value from the input voltage VIN atthe other end of the coil element 4008.

To regulate the value of the generated output voltage Vout to a givenvalue, the control semiconductor integrated circuit device 4007 controlsthe pulse width of the pulse width control signal PWM according to thevalue of the generated output voltage Vout.

<Structure of High-Side MOSFET and Low-Side MOSFET>

The structure of the high-side MOSFET 4005 and low-side MOSFET 4006 willthen be described. The high-side MOSFET 4005 and the low-side MOSFET4006 are different in size from each other but are identical instructure with each other. The structure of the low-side MOSFET 4006will be described as an example.

FIG. 37(A) is a diagrammatical plan view showing a layout of thelow-side MOSFET 4006 in a semiconductor chip. FIG. 37(B) is adiagrammatical sectional view showing a section of the low-side MOSFET4006 taken along a B-B′ line of FIG. 37(A). In FIG. 37(A), 3700 denotesa semiconductor chip. In this embodiment, two MOSFETs are formed in thesemiconductor chip 3700, and respective sources S, drains D, first gateelectrodes G1, and second gate electrodes G2 of the two MOSFETs areconnected to each other to make up a single low-side MOSFET 4006. Thestructure of the low-side MOSFET 4006, however, is not limited to thisstructure.

In FIG. 37(A), 3701 denotes respective source electrodes of the twoMOSFETs, 3702 denotes respective second gate electrodes G2 of the twoMOSFETs, and 3703 denotes a connection pad of the first gate electrodeG1 of the low-side MOSFET 4006. The drain electrode of the low-sideMOSFET 4006 is the back face of the semiconductor chip 3700. Each sourceelectrode 3701 is so formed as to cover a partial area of each secondgate electrode 3702. Using the area of second gate electrode 3702 thatis not covered with the source electrode 3701, the drive signal GL issupplied to the second gate electrode 3702.

In FIG. 37(B), 3704 denotes an n-type (first conductive) semiconductorregion that functions as the drain D of the low-side MOSFET 4006, 3705denotes a p-type (second conductive) semiconductor region in which thechannel of the MOSFET is formed, and 3706 denotes an n⁺-type (firstconductive) semiconductor region that functions as the source S of theMOSFET. The main surface of the semiconductor region 3704 is coveredwith the semiconductor region 3705 overlaid thereon, and the mainsurface of the semiconductor region 3705 is covered with thesemiconductor region 3706 overlaid thereon. As shown in FIG. 37(B),slots are formed in the semiconductor regions 3704, 3705, and 3706. In aslot formed in the semiconductor region 3704, a metal layer 3708 isformed to be adjacent to the semiconductor region 3704 across aninsulating layer 3707. In a slot formed in the semiconductor regions3704 and 3705, a metal layer 3709 is so formed as to overlap the metallayer 3708 and be adjacent to the semiconductor regions 3704 and 3705across the insulating layer 3707. The metal layer 3709 makes up thefirst gate electrode G1, while the metal layer 3708 makes up the secondgate electrode G2.

It is understood that the part of insulating layer 3707 that issandwiched between the metal layer 3709 functioning as the first gateelectrode G1 and the semiconductor region 3705 serves as a gateinsulating film for the low-side MOSFET 4006. According to thisembodiment, in the vertical direction of the slots, the metal layer 3709making up the first gate electrode G1 partially overlaps thesemiconductor region 3706 functioning as the source S and thesemiconductor region 3704 functioning as the drain D. In the verticaldirection of the slots, the metal layer 3708 functioning as the secondgate electrode G2 is embedded in the semiconductor region 3704functioning as the drain D. In other words, in the vertical direction ofthe slots, the second gate electrode G2 is located closer to the drain Dthan the first gate electrode G1.

In FIG. 37(B), 3701 denotes a source electrode connected electrically tothe semiconductor regions 3706 functioning as the source S. The sourceelectrode 3701 is connected electrically also to the semiconductorregions 3705 in which the channel is formed. Hence, the source S and theback gate of the low-side MOSFET 4006 are connected electrically to thesource electrode 3701. The metal layer 3709 functioning as the firstgate electrode G1 is connected to the connection pad 3703 shown in FIG.37(A). 3710 denotes a drain electrode formed on the back face of thesemiconductor chip 3700.

In FIG. 37(B), Crss denotes a first gate-drain capacitance formedbetween the first gate electrode G1 and the drain D. By determining avoltage supplied to the second gate electrode G2 to be a negativevoltage negative in polarity relative to the source S, a depletion layeris expanded significantly by the second gate electrode, which enables areduction in the first gate-drain capacitance Crss. By determining avoltage supplied to the second gate electrode G2 to be a positivevoltage positive in polarity relative to the source S, resistance in thedrain region corresponding to the second gate electrode G2 is reduced,which enables a reduction in the on-resistance of the low-side MOSFET4006 when it is switched on.

Similarly, in the case of the high-side MOSFET 4005, by changing thepolarity of a voltage supplied to the second gate electrode G2 (relativeto a voltage at the source as reference voltage), the first gate-draincapacitance Crss can be reduced and the on-resistance can also bereduced. The low-side MOSFET 4006 (FIG. 40) is constructed to be largerthan the high-side MOSFET 4005 because of a particular function of thelow-side MOSFET 4006 that it causes a current from the coil element 4008to flow to the ground voltage PGND node, thereby dropping a voltage atthe output terminal T1, which function will be described later referringto FIG. 36(B). Reducing the first gate-drain capacitance Crss andon-resistance of the low-side MOSFET 4006, therefore, is particularlyeffective for better efficiency.

First Embodiment

FIG. 1(A) is a circuit diagram showing a configuration of a principlepart of the semiconductor integrated circuit device 4002 according to afirst embodiment. FIG. 1(B) is a waveform chart showing the waveform ofa voltage in the semiconductor integrated circuit device 4002 of FIG.1(A).

FIG. 1(A) depicts the low-side MOSFET 4006 and the drive circuit 4012 inthe semiconductor integrated circuit device 4002 of FIG. 40. Componentsnot depicted in FIG. 1(A) are the same as the components of thesemiconductor integrated circuit device 4002 of FIG. 40 and aretherefore omitted in further description.

As described referring FIG. 40, the first gate electrode G1 of thelow-side MOSFET 4006 is connected to the output terminal T9 of thedriver 4003, the source and back gate of the same are connected to thevoltage terminal T2, and the drain of the same is connected to theoutput terminal T1 of the semiconductor integrated circuit device 4002.The driver 4003 has the drive circuit 4012 that drives the low-sideMOSFET 4006, and the drive signal GL from the drive circuit is suppliedto the first gate electrode G1 of the low-side MOSFET 4006 via theoutput terminal T9.

According to the first embodiment, the driver 4003 has a controlterminal T14 and a second gate electrode control circuit 1000 connectedto the control terminal T14. The control terminal T14 is connected tothe second gate electrode G2 of the low-side MOSFET 4006, so that asecond gate control signal UL generated by the second gate electrodecontrol circuit 1000 is supplied to the second gate electrode G2 of thelow-side MOSFET 4006 via the control terminal T14. The second gateelectrode control circuit 1000, for example, has a variable voltagesource 1001, as shown in FIG. 1(A). The variable voltage source 1001generates a positive voltage positive in polarity relative to the groundvoltage CGND, which positive voltage is variable in voltage value. Theground voltage CGND is the ground voltage GND substantially the same asthe ground voltage PGND. Hence, the second gate electrode controlcircuit 1000 generates the second gate control signal UL positive inpolarity relative to the source of the low-side MOSFET 4006 and having avariable voltage value.

Because a positive voltage positive in polarity relative to a voltage atthe source (ground voltage PGND) as a reference voltage is supplied asthe second gate control signal UL, to the second gate electrode G2 ofthe low-side MOSFET 4006, the on-resistance of the low-side MOSFET 4006when it is switched on can be reduced. Reducing the on-resistance allowsa reduction in the loss (power consumption) of the low-side MOSFET 4006when it is switched on, thus achieving a reduction in the loss of thesemiconductor integrated circuit device 4002. In the first embodiment,because a voltage supplied to the second gate electrode G2 is variablein voltage value, the on-resistance value can be adjusted.

Examples of the second gate electrode control circuit 1000 are shown inFIGS. 1(C) and 1(D). FIG. 1(C) is a circuit diagram showing an exampleof the second gate electrode control circuit 1000 that generates thesecond gate control signal UL having a positive voltage. FIG. 1(D) is acircuit diagram showing an example of the second gate electrode controlcircuit 1000 that generates the second gate control signal UL having anegative voltage. FIG. 1(B) is a waveform chart showing the voltagewaveform of the second gate control signal UL generated by the secondgate electrode control circuit 1000 of FIG. 1(D). In this embodiment,voltage polarity means polarity relative to the polarity of a voltage atthe source S of the low-side MOSFET 4006, i.e., the ground voltage GND.Therefore, a voltage with positive polarity means a positive voltage anda voltage with negative polarity means a negative voltage.

The second gate electrode control circuit 1000 that generates the secondgate control signal UL having a positive voltage will first bedescribed. In FIG. 1(C), 1002 denotes an n-channel type MOSFET, 1003 and1004 denote resistance elements, 1005 denotes a differential amplifiercircuit, and 1006 denotes a variable voltage source.

The resistance elements 1003 and 1004 are connected between theconnection terminal T14 and the ground voltage CGND node, and a dividedvoltage is extracted from a connection node between the resistanceelement 1003 and the resistance element 1004. The extracted dividedvoltage is supplied to the reverse input terminal (−) of thedifferential amplifier circuit 1005, while a variable voltage from thevariable voltage source 1006 is supplied to the non-reverse inputterminal (+) of the differential amplifier circuit 1005. An outputsignal from the differential amplifier circuit 1005 is supplied to thegate of the MOSFET 1002 whose drain is supplied with the source voltageVCIN and whose back gate and drain are connected to the control terminalT14. The differential amplifier circuit 1005 controls the MOSFET 1002 sothat a voltage difference between the divided voltage determined by aresistance ratio between the resistance element 1003 and the resistanceelement 1004 and the variable voltage from the variable voltage source1006 is reduced. As a result, a voltage corresponding to the variablevoltage from the variable voltage source 1006 is generated as the secondgate control signal UL and is supplied to the second gate electrode G2of the low-side MOSFET 4006. In this example, the on-resistance value ofthe low-side MOSFET 4006 can be adjusted by changing the variablevoltage value of the variable voltage source 1006.

The second gate electrode control circuit 1000 that generates the secondgate control signal UL having a negative voltage will then be described,referring to FIGS. 1(D) and 1(B). In FIG. 1(D), the second gateelectrode control circuit 1000 has a p-channel type MOSFET 1007, ann-channel type MOSFET 1008, an oscillation circuit 1013, capacitorelements 1009 and 1012, and diode elements 1010 and 1011.

The p-channel type MOSFET 1007 and the n-channel type MOSFET 1008 havetheir respective source-drain paths connected in series between thesource voltage VCIN node and the ground voltage CGND node. Anoscillation output signal from the oscillation circuit 1013 is suppliedto respective gate electrodes of the p-channel type MOSFET 1007 and then-channel type MOSFET 1008. In other words, the p-channel type MOSFET1007 and the n-channel type MOSFET 1008 make up a CMOS-type inverter, towhich the oscillation output signal from the oscillation circuit 1013 isinput. The output end of the inverter (connection node between theMOSFET 1007 and the MOSFET 1008) is connected to the cathode of thediode element 1010 and the anode of the diode element 1011 via thecapacitor element 1009, and the anode of the diode element 1010 isconnected to one end of the capacitor element 1012 and to the controlterminal T14. The cathode of the diode element 1011 and the other end ofthe capacitor element 1012 are connected to the ground voltage CGNDnode.

According to the oscillation output signal from the oscillation circuit1013, the inverter (MOSFETs 1007 and 1008) causes the capacitor element1009 to discharge and be charged cyclically. When the MOSFET 1007 isswitched on, the MOSFET 1007, the capacitor element 1009, and the diodeelement 1011 jointly form a charge path through which the capacitorelement 1009 is charged. When the MOSFET 1008 is switched on, on theother hand, the MOSFET 1008, the capacitor element 1009, the diodeelement 1010, and the capacitor element 1012 jointly forma dischargepath. When the discharge path is formed, electric charges aredistributed between the capacitor element 1009 and the capacitor element1012, as a result of which a voltage at the control terminal T14 becomesa negative voltage (voltage with negative polarity) negative in polarityrelative to the ground voltage CGN. This negative voltage is supplied asthe second gate control signal UL, to the second gate electrode G2 ofthe low-side MOSFET 4006.

FIG. 1(B) depicts a voltage at the source of the low-side MOSFET 4006(source voltage (GND) in FIG. 1(B)) and the voltage waveform of thesecond gate control signal UL. In FIG. 1(B), the horizontal axisrepresents time and the vertical axis represents voltage. The source Sand back gate of the low-side MOSFET 4006 are connected to the voltageterminal T2 supplied with the ground voltage PGND. A voltage at thesource of the low-side MOSFET 4006, therefore, matches the groundvoltage PGND (which is depicted as source voltage (GND) in FIG. 1(B)).The second gate electrode control circuit 1000 of FIG. 1(D) generates anegative voltage negative in polarity relative to the ground voltageCGND. Since both ground voltage PGND and ground voltage CGND are theground voltage (GND), the second gate control signal UL generated by thesecond gate electrode control circuit 1000 of FIG. 1(D) has a voltagelower than the voltage at the source of the low-side MOSFET 4006. Inother words, the voltage of the second gate control signal UL is anegative voltage with respect to the voltage at the source S defined asa reference voltage.

By supplying a negative voltage negative in polarity relative to avoltage at the source S of the low-side MOSFET 4006, to the second gateG2 of the low-side MOSFET 4006, the first gate-drain capacitance Crsscan be reduced. As a result, an on-off shift time (which willhereinafter be also referred to as “on-off transition time”) which thelow-side MOSFET 4006 takes to shift from its on-state to off-state orvice versa can be shortened. Shortening the on-off transition timeduring which power is consumed leads to a reduction in power consumption(loss) by the low-side MOSFET 4006, thus achieving a reduction in theloss of the semiconductor integrated circuit device 4002.

Second Embodiment

According to the first embodiment, the second gate electrode controlcircuit 1000 generates the second gate control signal UL having apositive voltage or negative voltage positive or negative in polarityrelative to a voltage at the source S. In the first embodiment, thesecond gate control signal UL having a positive or negative voltage issupplied constantly to the second gate electrode G2 in a case where thelow-side MOSFET 4006 is in a state of on-off transition caused by thedrive signal GL supplied to its first gate electrode G1 and also in acase where the low-side MOSFET 4006 is in an on-state or off-state.

However, examinations by the inventor of the present invention hasrevealed that when the second gate control signal UL having a negativevoltage is supplied to the second gate electrode G2, the firstgate-drain capacitance Crss reduces but the on-resistance of the MOSFETincreases. Likewise, when the second gate control signal UL having apositive voltage is supplied to the second gate electrode G2, theon-resistance of the MOSFET reduces but the first gate-drain capacitanceCrss increases. This means that constantly supplying the second gatecontrol signal UL having a positive or negative voltage to the secondgate electrode G2 of the MOSFET results in a loss increase in somecases.

To solve this problem, the inventor of the present invention has studiedabout loss caused by the on-resistance of the MOSFET, i.e., conductionloss and loss caused when the MOSFET in its on-state is switched off orthe MOSFET in its off-state is switched on, i.e., switching loss. Thesemiconductor integrated circuit device 4002 used in the power supplysystem 4000 has been studied to examine the types and ratios of lossesthe semiconductor integrated circuit device 4002 suffers.

Losses the semiconductor integrated circuit device 4002 suffers havebeen classified. FIG. 38 is a characteristics diagram showing the lossof the semiconductor integrated circuit device 4002. Three measurementresults are shown in FIG. 38, at the center of which a characteristicsgraph is indicated, which represents the relation between an outputcurrent (load current) Iout (A) flowing through the output terminal T1(FIG. 40) and the efficiency (%) of the semiconductor integrated circuitdevice 4002. In this characteristics graph, the horizontal axisrepresents output current and the vertical axis represents a ratiobetween the input power and output power (output power/input powerratio) of the semiconductor integrated circuit device 4002. As the load4001 of the power supply system 4000 increases in size and growsheavier, the load current (output current) Iout grows larger. This isbecause that the heavier load requires a higher current.

It is understood from the characteristics diagram of FIG. 38 that theefficiency is high when the load current Iout is relatively low and thatthe efficiency gets lower as the load current Iout gets higher. A casewhere the value of the load current (output current) Iout is equal to orsmaller than a given current value i2 is considered to be a light loadcase, and a case where the value of the load current (output current)Iout is larger than the given current value i2 is considered to be aheavy load case. Based on this assumption, the types and ratios oflosses are determined for the light load case and heavy load case. Onthe left side in FIG. 38, the types and ratios of losses in the lightload case where the load current (output current) Iout takes a currentvalue i1 equal to or smaller than the given current value i2 areindicated as “loss breakdown in the light load case”. In the samemanner, on the right side in FIG. 38, the types and ratios of losses inthe heavy load case where the load current (output current) Iout takes acurrent value i3 larger than the given current value i2 are indicated as“loss breakdown in the heavy load case”.

Each of “loss breakdown in the light load case” and “loss breakdown inthe heavy load case” expresses loss items in the form of stacked bars.Stacked bars of items represent the types of losses, i.e., switchingloss (hereinafter, also referred to as “SW loss”), conduction loss, andother loss. The types of losses will then be described.

Other loss represents a loss caused by a logical circuit in thesemiconductor integrated circuit device 4002, e.g., a loss caused by thedriver 4003. SW loss and conduction loss represents losses caused by thehigh-side MOSFET 4005 and low-side MOSFET 4006, which will be describedreferring to FIGS. 39(A) and 39(B) that are explanatory diagrams of aswitching loss and a conduction loss.

A figure on the upper side in FIG. 39(A) is a diagrammatical view of achange in a source-drain voltage VDS, a change in a drain current IDS,and a loss P that arises when the low-side MOSFET 4006 (high-side MOSFET4005) shifts from its off-state to on-state. A figure on the lower sidein FIG. 39(A) is a characteristics graph showing the relation betweenthe voltage of the second gate control signal UL supplied to the secondgate electrode G2 and the first gate-drain capacitance Crss. In thischaracteristics graph, the horizontal axis represents the voltage of thesecond gate control signal UL (UL voltage in FIG. 39(A)) and thevertical axis represents the value of the first gate-drain capacitanceCrss.

The SW loss, i.e., switching loss P is the loss that arises when thelow-side MOSFET (high-side MOSFET) shifts from its off-state to on-state(or from on-state to off-state). During on-off shift, as indicated bythe figure on the uppers side in FIG. 39(A), the source-drain voltageVDS takes a definite value in a certain period and the drain current IDStakes a definite value in a certain period and both periods overlap in acertain timespan. Power is consumed during this timespan, which isregarded as the switching loss P. The switching loss P is, therefore,proportional to the product of the voltage (VDS) and the current (IDS).A time required for on-off shift, on the other hand, depends on acapacitance accompanying the first gate electrode of the MOSFET. Thefirst gate-drain capacitance Crss is such a capacitance accompanying thefirst gate electrode of the MOSFET. As indicated by the characteristicsgraph on the lower side in FIG. 39(A), the first gate-drain capacitanceCrss can be reduced by lowering the voltage of the second gate controlsignal UL supplied to the second gate electrode G2 to give the voltagenegative polarity.

Reducing the first gate-drain capacitance Crss hastens the change in thesource-drain voltage VDS and drain current IDS, thereby shortens thetime required for on-off shift. As a result, the switching loss P isreduced.

As indicated by a formula on the upper side in FIG. 39(B), theconduction loss is the loss proportional to the product of theon-resistance (Ron) of the low-side MOSFET (high-side MOSFET) and thesquare of the drain current IDS (IDS²). On the lower side in FIG. 39(B),a characteristics graph is indicated as a graph representing therelation between the voltage of the second gate control signal UL (ULvoltage) and the on-resistance. In this characteristics graph, thehorizontal axis represents the voltage of the second gate control signalUL and the vertical axis represents the on-resistance value of thelow-side MOSFET (high-side MOSFET). It is understood from thecharacteristics graph on the lower side in FIG. 39(B) that theon-resistance Ron of the low-side MOSFET (high-side MOSFET) is reducedby causing the voltage of the second gate control signal UL (UL voltage)supplied to the second gate electrode G2 to shift from a negativevoltage to a positive voltage.

The loss breakdown in the light load case and the loss breakdown in theheavy load case shown in FIG. 38 will then be described. It isunderstood from the “breakdown of loss with light load” on the left sidein FIG. 38 that in the light load case, the ratio of the “SW loss” ishigher than the ratio of the “conduction loss” and “other loss”. Incontrast, it is understood from the “breakdown of loss with heavy load”on the right side in FIG. 38 that in the heavy load case, the ratio ofthe “conduction loss” is higher than the ratio of the “SW loss” and“other”, which indicates that as the load becomes heavier, the ratio ofthe “conduction loss” to the total loss of the semiconductor integratedcircuit device becomes higher. In other words, the ratio of theconduction loss and the ratio of the switching loss to the total loss ofthe semiconductor integrated circuit device vary depending on the load.When the load is heavy, the ratio of the conduction loss is high. Whenthe load is light, the ratio of the switching loss is high.

In a semiconductor integrated circuit device according to a secondembodiment to be described next, both switching loss and conduction lossare reduced.

FIG. 2(A) is a block diagram showing a configuration of a principle partof the driver 4003 in the semiconductor integrated circuit device 4002according to a second embodiment. The second gate electrode controlcircuit 1000 is depicted as the principle part. In the secondembodiment, the configuration of the second gate electrode controlcircuit 1000 shown in FIG. 1(A) is modified to the configuration shownin FIG. 2(A). FIG. 2(A) depicts the control terminal T14 of the driver4003 and the second gate electrode control circuit 1000 but does notdepict the low-side MOSFET 4006 and the drive circuit 4012 that areshown in FIG. 1(A). The second gate electrode control circuit 1000 ofFIG. 2(A) is disposed in the driver 4003 of FIG. 40.

In FIG. 2(A), the second gate electrode control circuit 1000 has apositive voltage regulator 2000, a negative voltage regulator 2001, alevel shifter 2003, a selecting circuit 2002, and a second gateelectrode drive control circuit 2004. The positive voltage regulator2000 generates a positive voltage Vpos positive in polarity relative tothe ground voltage PGND, and the negative voltage regulator 2001generates a negative voltage Vneg negative in polarity relative to theground voltage PGND. The generated positive voltage Vpos and negativevoltage Vneg are supplied to the level shifter 2003 and to the selectingcircuit 2002.

The second gate electrode drive control circuit 2004 receives the drivesignal GL output from the drive circuit 4012 (FIG. 40), generates acontrol signal synchronizing with the drive signal GL, and supplies thegenerated control signal to the level shifter 2003. Receiving thegenerated control signal from the second gate electrode drive controlcircuit 2004, the level shifter 2003 shifts the high-voltage level andthe low-voltage level of the control signal to a voltage level matchingthe positive voltage Vpos and a voltage level matching the negativevoltage Vneg, respectively, and supplies the control signal with shiftedvoltage levels to the selecting circuit 2002.

According to the voltage of the supplied control signal (high-voltagelevel/low-voltage level), the selecting circuit 2002 selects either thepositive voltage Vpos or negative voltage Vneg, and outputs a selectedvoltage (positive voltage Vpos or negative voltage Vneg) as the secondgate control signal UL, to the terminal T14. As shown in FIG. 1(A), theterminal T14 is connected to the second gate electrode G2 of thelow-side MOSFET 4006. The control signal supplied from the level shifter2003 to the selecting circuit 2002 synchronizes with the drive signal GLoutput from the drive circuit 4012. Hence, the voltage of the secondgate control signal UL supplied to the second gate electrode G2 of thelow-side MOSFET 4006 synchronizes with the voltage of the drive signalGL that switches on and off the low-side MOSFET 4006, thus matching thenegative voltage Vneg or positive voltage Vpos.

FIG. 2(B) depicts a circuit configuration example of the selectingcircuit 2002, the level shifter 2003, and the second gate electrodedrive control circuit 2004. The second gate electrode drive controlcircuit 2004 has three inverters connected in parallel between thesupply voltage VCIN node and the ground voltage CGND node, whichinverters are CMOS type inverters composed of p-channel type MOSFETs2009, 2008, and 2007 and n-channel type MOSFETs 2015, 2014, and 2013,respectively. The inverters are cascaded such that the input end of eachof the second and third inverters is connected to the output end of eachof the first and second inverters. The first inverter (composed of thep-channel type MOSFET 2009 and the n-channel type MOSFET 2015) issupplied with the drive signal GL from the drive circuit 4012 (FIG. 40),and an output signal from the third inverter is supplied to the levelshifter 2003. Hence, the second gate electrode drive control circuit2004 of FIG. 2(B) supplies a signal created by reversing the phase ofthe drive signal GL as a control signal, to the level shifter 2003.

The level shifter 2003 has an n-channel type MOSFET 2012 that receivesthe control signal from the second gate electrode drive control circuit2004, and a load element 2016. The drain of the n-channel type MOSFET2012 is connected to the positive voltage Vpos node via the load element2016. In the example of FIG. 2(B), the source of the n-channel typeMOSFET 2012 is connected to the ground voltage CGND node. As a result, acontrol signal that shifts in voltage between the positive voltage Vposand the ground voltage CGND is output from a connection node between theload element 2016 and the n-channel type MOSFET 2012, which means thatthe control signal shifting in voltage is output from the level shifter2003. In FIG. 2(B), the source of the n-channel type MOSFET 2012 isconnected to the ground voltage CGND node. The source, however, may beconnected to the negative voltage Vneg node.

The selecting circuit 2002 has two inverters connected in parallelbetween the positive voltage Vpos node and the negative voltage Vnegnode, which inverters are CMOS type inverters composed of n-channel typeMOSFETs 2011 and 2010 and p-channel type MOSFETs 2006 and 2005,respectively. The inverters are cascaded such that the input end of thesecond inverter is connected to the output end of the first inverter.The input end of the first inverter (composed of the n-channel typeMOSFET 2011 and the p-channel type MOSFET 2006) is supplied with thecontrol signal from the level shifter 2003, and the output end of thesecond inverter (composed of the n-channel type MOSFET 2010 and thep-channel type MOSFET 2005) is connected to the control terminal T14.Each inverter in the selecting circuit 2002 operates with the positivevoltage Vpos and negative voltage Vneg serving as supply voltages.Hence, the second inverter selects either the positive voltage Vpos ornegative voltage Vneg according to the control signal from the levelshifter 2003 and outputs the selected voltage to the control terminalT14.

Various configurations of the positive voltage regulator 2000 and thenegative voltage regulator 2001 are possible. For example, the circuitsshown in FIGS. 1(C) and 1(D) may be adopted as configurations of thepositive voltage regulator 2000 and the negative voltage regulator 2001.

FIGS. 3(A) to 3(D) are operation waveform charts of the semiconductorintegrated circuit device 4002 having the second gate electrode controlcircuit 1000 of FIGS. 2(A) and 2(B). The operation of the semiconductorintegrated circuit device 4002 according to the second embodiment willthen be described, referring to FIGS. 1(A), 2(A), FIGS. 3(A) to 3(D),and FIG. 40.

In FIGS. 3(A) to 3(D), the horizontal axis and the vertical axisrepresent time and voltage, respectively, and a period (a) represents aperiod during which the high-side MOSFET 4005 is on and the low-sideMOSFET 4006 is off while a period (b) represents a period during whichthe high-side MOSFET 4005 is off and the low-side MOSFET 4006 is on. Asdescribed above referring to FIGS. 40 and 41, the high-side MOSFET 4005and the low-side MOSFET 4006 are switched on and off complementally bythe drive signals GH and GL.

FIG. 3(A) depicts the waveform of the output voltage VSWH at the outputterminal T1 of the semiconductor integrated circuit device 4002, FIG.3(B) depicts the waveform of the drive signal GH from the drive circuit4011 (FIG. 40), and FIG. 3(C) depicts the waveform of the drive signalGL from the drive circuit 4012 (FIG. 40). FIG. 3(D) depicts the waveformof the second gate control signal UL output from the second gateelectrode control circuit 1000 of FIG. 2(A).

It is understood from the description of FIGS. 2(A) and 2(B) that thevoltage of the second gate control signal UL synchronizes with thevoltage of the drive signal GL, thus shifting between the positivevoltage Vpos and the negative voltage Vneg. The source S of the low-sideMOSFET 4006 is connected to the voltage terminal T2 whose potential isthe ground voltage (FIG. 1(A) and FIG. 40). The voltage of the secondgate control signal UL, therefore, matches the positive voltage(positive voltage Vpos) or negative voltage (negative voltage Vneg)positive or negative in polarity relative to a voltage at the source S(source voltage (GND)) of the low-side MOSFET 4006 serving as areference voltage. According to the second embodiment, as indicated inFIG. 2(B), when the drive signal GL supplied to the first gate electrodeG1 of the low-side MOSFET 4006 shifts to a high-voltage level signal,the second gate electrode control circuit 1000 outputs the positivevoltage Vpos as the second gate control signal UL, in synchronizationwith the voltage level shift of the drive signal GL. When the drivesignal GL shifts to a low-voltage level signal, the second gateelectrode control circuit 1000 outputs the negative voltage Vneg as thesecond gate control signal UL, in synchronization with the voltage levelshift of the drive signal GL.

At time t1, the drive signal GL output from the drive circuit 4012changes in voltage level from a high-voltage level to a low-voltagelevel. Because this drive signal GL is supplied to the first gateelectrode G1 of the low-side MOSFET 4006, the low-side MOSFET 4006shifts from its on-state to off-state. In response to the shift of thedrive signal GL to its low-voltage level, the voltage of the second gatecontrol signal UL output from the second gate electrode control circuit1000 shifts to the negative voltage Vneg at time t1. Because the secondgate control signal UL is supplied to the second gate electrode G2 ofthe low-side MOSFET 4006, the shift of the second gate control signal ULto the negative voltage Vneg results in a reduction in the firstgate-drain capacitance Crss at the low-side MOSFET 4006. As a result,the low-side MOSFET 4006 shifts from its on-state to off-state morequickly, thus shortening its on-off shift time.

To prevent the high-side MOSFET 4005 and low-side MOSFET 4006 fromswitching on simultaneously, the drive signal GH output from the drivecircuit 4011 changes in voltage level from a high-voltage level to alow-voltage level at time t2 after an elapse of a given time (dead timeperiod) from time t1. Because the drive signal GH is supplied to thefirst gate electrode G1 of the high-side MOSFET 4005, the high-sideMOSFET 4005 shifts from its off-state to on-state. This raises theoutput voltage VSWH at the output terminal T1.

Although both high-side MOSFET 4005 and low-side MOSFET 4006 are offduring the dead time period (between time t1 and time t2), the voltageVSWH at the output terminal T14 drops. This is partly caused byswitching loss that arises in a period during which the low-side MOSFET4006 shifts from its on-state to off-state. According to the secondembodiment, the on-off shift period can be shortened, so that theswitching loss can be reduced.

Subsequently, at time t3, the drive signal GH at its high-voltage levelshifts to a low-voltage level. As a result, the high-side MOSFET 4005shifts from its on-state to off-state. At time t4 after an elapse of thetime equivalent to the dead time period from time t3, the drive signalGL at its low-voltage level shifts to a high-voltage level. This voltagelevel shift of the drive signal GL causes the low-side MOSFET 4006 toshift from its off-state to on-state and also causes the second gateelectrode control circuit 1000 to shift the voltage of the second gateelectrode control signal UL output from the second gate electrodecontrol circuit 1000 to the positive voltage Vpos.

At this point, because the high-side MOSFET 4005 is off while thelow-side MOSFET 4006 is on, the output voltage VSWH at the outputterminal T14 drops. When the high-voltage level drive signal GL keepsthe low-side MOSFET 4006 on, the second gate control signal UL havingthe positive voltage Vpos is supplied to the second gate electrode G2 ofthe low-side MOSFET 4006. This reduces the on-resistance of the low-sideMOSFET 4006, thereby reduces the conduction loss of the low-side MOSFET4006.

At time t5, the drive signal GL at its high-voltage level shifts to alow-voltage level again. Afterward, the above operations at time t1 tot4 are repeated.

In the second embodiment, when the low-side MOSFET 4006 is switched onby the drive signal GL supplied to the first gate electrode G1, thepositive voltage Vpos is supplied from the second gate electrode controlcircuit 1000 that operates according to the drive signal GL, to thesecond gate electrode G2. When the low-side MOSFET 4006 is switched offby the drive signal GL supplied to the first gate electrode G1, thenegative voltage Vneg is supplied from the second gate electrode controlcircuit 1000 that operates according to the drive signal GL, to thesecond gate electrode G2. Hence, when the low-side MOSFET 4006 isswitched on, the conduction loss caused by the on-resistance of theMOSFET is reduced. When the low-side MOSFET 4006 is switched between itson-state and off-state (on-state and off-state), the switching loss isreduced.

FIGS. 4(A) and 4(D) are explanatory diagrams of a reduction in theconduction loss. FIG. 4(D) depicts the circuit of the low-side MOSFET4006. The low-side MOSFET 4006 has its source S connected to the groundvoltage PGND (GND) node and its drain D connected to the controlterminal T14. In FIG. 4(D), Ron denotes the on-resistance of the MOSFET4006 that results when it is on, and ISD denotes a source-drain currentflowing through the MOSFET 4006 in its on-state. It is assumed that whenthe low-side MOSFET 4006 is switched on, a current is supplied from theground voltage node to the one end of the coil element 4008 (FIG. 40)via the low-side MOSFET 4006. In FIG. 4(D), therefore, the suppliedcurrent is not indicated as the drain current IDS but as thesource-drain current ISD.

FIG. 4(A), similar to FIG. 3(A), depicts the waveform of the outputvoltage VSWH at the output terminal T1. FIG. 4(B) is an enlarged view ofFIG. 4(A), showing an enlarged view of the waveform of the outputvoltage VSWH in a period during which the low-side MOSFET 4006 remainson. FIG. 4(C) depicts the waveform of the source-drain current ISD inthe period during which the low-side MOSFET 4006 remains on.

In FIG. 4(B), a broken line indicates the waveform of the output voltageVSWH that results when a negative voltage is applied to the second gateelectrode G2 of the low-side MOSFET 4006, and a continuous lineindicates the waveform of the output voltage VSWH that results when apositive voltage is applied to the second gate electrode G2 of thelow-side MOSFET 4006. Applying a negative voltage to the second gateelectrode G2 increases the on-resistance. In contrast, according to thesecond embodiment, applying a positive voltage to the second gateelectrode G2 reduces the on-resistance. It is known that anon-resistance voltage is given as the product of a resistance (Ron) anda current (source-drain current ISD). By reducing the on-resistance,therefore, the conduction loss is reduced. Reducing the on-resistancealso prevents a case where the output voltage VSWH drops excessivelywhen the low-side MOSFET is switched on.

Third Embodiment

FIG. 5 is a block diagram showing a configuration of the second gateelectrode control circuit 1000 in the semiconductor integrated circuitdevice 4002 according to a third embodiment. To the third embodiment,knowledge based on the study by the inventor described in the secondembodiment is applied. The ratios of the “conduction loss” and“switching loss” of the semiconductor integrated circuit device varydepending on a load such that the ratio of the “conduction loss” becomeshigher as the load becomes heavier. Based on this knowledge, a loss witha high ratio is reduced so that the total loss of the semiconductorintegrated circuit device is reduced efficiently.

In the third embodiment, in the same manner as in the first and secondembodiments, the driver 4003 (FIG. 40) has the second gate electrodecontrol circuit 1000 and the control terminal T14. In the same manner asin the second embodiment, the control terminal T14 is connected to thesecond gate electrode G2 of the low-side MOSFET 4006 (FIGS. 1 and 40).Other components and operations of the driver 4003 are the same as thosedescribed in FIG. 40 and are therefore omitted in further description.

In FIG. 5, the second gate electrode control circuit 1000 has a loadcurrent detecting circuit 5000, a second gate electrode drive controlcircuit 5001, a positive voltage regulator 5002, a negative voltageregulator 5003, and switches 5004 and 5005. In the same manner as in thesecond embodiment, the second gate electrode control circuit 1000generates the second gate electrode control signal UL having a positivevoltage or negative voltage positive or negative in polarity relative tothe ground voltage GND as a reference voltage, and supplies the secondgate electrode control signal UL to the second gate electrode G2 of thelow-side MOSFET 4006 via the control terminal T14.

The positive voltage regulator 5002 receives a control signal 5006output from the second gate electrode drive control circuit 5001, as anon/off signal that turns on and off the positive voltage regulator 5002.When the control signal 5006 instructs to turn on the positive voltageregulator 5002, the positive voltage regulator 5002 generates thepositive voltage Vpos positive in polarity relative to the groundvoltage. When the control signal 5006 instructs to turn off the positivevoltage regulator 5002, the negative voltage regulator 5002 is turnedoff.

Similar to the positive voltage regulator 5002, the negative voltageregulator 5003 receives a control signal 5007 output from the secondgate electrode drive control circuit 5001, as an on/off signal thatturns on and off the negative voltage regulator 5003. When the controlsignal 5007 instructs to turn on the negative voltage regulator 5003,the negative voltage regulator 5003 generates the negative voltage Vnegnegative in polarity relative to the ground voltage. When the controlsignal 5007 instructs to turn off the negative voltage regulator 5003,the negative voltage regulator 5003 is turned off.

The switch 5004 is switched on and off according to the control signal5006. When the switch 5004 is switched on, the positive voltage Vposgenerated by the positive voltage regulator 5002 is supplied to thecontrol terminal T14. The switching on/off of the switch 5004synchronizes with turning on/off of the positive voltage regulator 5002.This means that when the positive voltage regulator 5002 is turned on tooperate by the control signal 5006, the switch 5004 is in its on-stateand that when the positive voltage regulator 5002 is turned off to stopoperating by the control signal 5006, the switch 5004 is in itsoff-state.

Similar to the switch 5004, the switch 5005 is switched on and offaccording to the control signal 5007. When the switch 5005 is switchedon, the negative voltage Vneg generated by the negative voltageregulator 5003 is supplied to the control terminal T14. The switchingon/off of the switch 5005 synchronizes with turning on/off of thenegative voltage regulator 5003. This means that when the negativevoltage regulator 5003 is turned on to operate by the control signal5007, the switch 5005 is in its on-state and that when the negativevoltage regulator 5003 is turned off to stop operating by the controlsignal 5007, the switch 5005 is in its off-state.

The second gate electrode drive control circuit 5001 receives adetection signal from the load current detecting circuit 5000, andgenerates the control signal 5006 and the control signal 5007 accordingto, for example, the voltage of the detection signal, thereby turns onor off the positive voltage regulator 5002 or negative voltage operator5003. The second gate electrode drive control circuit 5001 also switcheson the switch (switch 5004 or switch 5005) corresponding to theregulator turned on (positive regulator 5002 or negative regulator5003). Through this process, according to the detection signal from theload current detecting circuit 5000, the second gate electrode controlcircuit 1000 outputs the second gate electrode control signal UL havingthe positive voltage Vpos or negative voltage Vneg, to the terminal T14.

FIGS. 6(A) and 6(B) are waveform charts showing the operation of thesecond gate electrode control circuit 1000 of FIG. 5. In FIGS. 6(A) and6(B), the horizontal axis represents time. FIG. 6(A) depicts thewaveform of the load current Iout flowing through the output terminal T1of the semiconductor integrated circuit device 4002, and the verticalaxis of FIG. 6(A) represents current value. FIG. 6(B) depicts thewaveform of the second gate electrode control signal UL output from thesecond gate electrode control circuit 1000 of FIG. 5, and the verticalaxis of FIG. 6(B) represents voltage values.

As described referring to FIG. 38, the value of the load current Ioutchanges depending on whether the load 4001 connected to the outputterminal T13 of the power supply system 4000 (FIG. 40) is heavy orlight. To put it another way, the value of the load current Iout growslarger as the load becomes heavier.

Receiving the load current Iout flowing through the output terminal T1,the load current detecting circuit 5000 of FIG. 5 generates a detectionsignal indicating whether the value of the load current Iout exceeds agiven current value (current i2 in FIG. 38) and supplies the generateddetection signal to the second gate electrode drive control circuit 5001(the operation of the load current detecting circuit 5000 is not limitedto this particular action). When the detection signal indicates that thevalue of the load current Iout exceeds the given current value, thesecond gate electrode drive control circuit 5001 turns the positiveregulator 5002 on and switches the switch 5004 on, while turns thenegative regulator 5003 off and the switches the switch 5005 off. Hence,when the load current Iout exceeds the given current value, the secondgate control signal UL having the positive voltage Vpos is supplied tothe second gate electrode G2 of the low-side MOSFET 4005.

When the value of the load current Iout is equal to or smaller than thegiven current value, the second gate electrode drive control circuit5001 turns the positive regulator 5002 off and switches the switch 5004off, while turns the negative regulator 5003 on and the switches theswitch 5005 on. Hence, when the load current Iout is equal to or smallerthan the given current value, the second gate control signal UL havingthe negative voltage Vneg is supplied to the second gate electrode G2 ofthe low-side MOSFET 4005.

The switches 5004 and 5005, therefore, can be regarded as selectingcircuits that output the positive voltage Vpos and the negative voltageVneg as the second gate control signals UL, respectively, according tothe detection signal from the load current detecting circuit 5000.

The third embodiment will be described referring to FIGS. 6(A) and 6(B).In a period (a), the value of the load current Iout remains equal to orsmaller than the given current value (e.g., current i2 in FIG. 38). Forthis reason, the second gate control signal UL output from the secondgate electrode control circuit 1000 has the negative voltage Vneg. In aperiod (b), in contrast, the value of the load current Iout remainslarger than the given current value. For this reason, the second gatecontrol signal UL output from the second gate electrode control circuit1000 has the positive voltage Vpos.

The load current Iout is low when the load is light, and is high whenthe load is heavy. According the third embodiment, when the load islight (period (a)), the negative voltage Vneg is supplied to the secondgate electrode G2 of the low-side MOSFET, which reduces the switchingloss. When the load is heavy (period (b)), the positive voltage Vpos issupplied to the second gate electrode G2 of the low-side MOSFET, inwhich case, therefore, the conduction loss of the low-side MOSFET isreduced. As described referring to FIG. 38, the ratio of the conductionloss is high when the load is heavy while the ratio of the switchingloss is high when the load is light. According to the third embodiment,when the load is heavy, the conduction loss at its high ratio in theheavy load case is reduced, and when the load is light, the switchingloss at its high ratio in the light load case is reduced. In thismanner, proper loss reduction can be performed according to thecondition of the load.

Various configurations of the load current detecting circuit 5000 arepossible.

Fourth Embodiment

FIG. 7 is a block diagram showing a configuration of the semiconductorintegrated circuit device 4002 according to a fourth embodiment. To thefourth embodiment, knowledge based on the study by the inventordescribed in the second embodiment is applied.

According to the fourth embodiment, the driver 4003 described referringto FIG. 40 further includes the control terminal T14, a load currentdetecting comparator 7000, a four-cycle detecting circuit 7001, ananalog switch 7003, an inverter 7002, the positive regulator 2000, andthe negative regulator 2001. The positive regulator 2000 and thenegative regulator 2001 have been described in FIG. 2(A) and aretherefore omitted in further description.

As shown in FIG. 40, the driver 4003 has multiple terminals. FIG. 7depicts the terminals described in FIG. 40 (voltage terminals T8 and T10and output terminal T9) out of the multiple terminals. As described inthe above multiple embodiments, the driver 4003 has the control terminalT14. The first gate electrode G1 of the low-side MOSFET 4006 isconnected to the output terminal T9, through which the drive signal GLfrom the drive circuit 4012 is supplied to the first gate electrode G1.The voltage terminal T10 is connected to the source S of the low-sideMOSFET 4006 and to the ground voltage PGND node. The voltage terminal T8is connected to the drain D of the low-side MOSFET 4006.

In the fourth embodiment, the voltage terminals T8 and T10 are connectedto the load current detecting comparator 7000. As shown in FIG. 40, thevoltage terminal T8 is connected also to the output terminal T1 of thesemiconductor integrated circuit device 4002. The voltage VSWH at thevoltage terminal T8, therefore, changes depending on an output signalfrom the semiconductor integrated circuit device 4002.

The load current detecting comparator 7000 includes a comparator 7004having a reverse input terminal (−) and a non-reverse input terminal(+), and an offset circuit 7005. The non-reverse input terminal (+) ofthe comparator 7004 is connected to the voltage terminal T10 and thereverse input terminal (−) of the same is connected to the voltageterminal T8 via the off-set circuit 7005. Because various configurationsof the offset circuit 7005 are possible, the offset circuit 7005 isindicated by a battery symbol in FIG. 7. When the low-side MOSFET 4006is off, the load current detecting comparator 7000 compares the voltagePGND at the voltage terminal T10 with the voltage VSWH at the voltageterminal T8.

When the low-side MOSFET 4006 is switched on by the drive signal GLsupplied from the drive circuit 4012 to the first gate electrode G1 viathe output terminal T9, the current ISD is supplied from the groundvoltage PGND node to the one end of the coil element 4018 (FIG. 40).This flow of the current ISD causes a voltage drop, thus causing thevoltage VSWH at the voltage terminal T8 to drop. As a result, a voltagegiven by adding an offset voltage created by the off-set circuit 7005 tothe dropped voltage (VSWH) at the voltage terminal T8 (hereinafter“voltage VSWH+offset) is supplied to the reverse input terminal (−) ofthe comparator 7004. Meanwhile, the ground voltage PGND (GND) issupplied to the non-reverse input terminal (+) of the comparator 7004.The comparator 7004 thus determines whether the voltage VSWH+offset ishigh or low relative to the ground voltage PGND, generates an outputsignal having a high-level voltage or low-level voltage as the result ofthe determination, and sends out the generated output signal as theoutput signal from the load current detecting comparator 7000.

The output signal from the load current detecting comparator 7000 issupplied to the four-cycle detecting circuit 7001. The four-cycledetecting circuit 7001 has a counter 7007 and an RS-type flip-flop 7006.The counter 7007 counts output signals from the load current detectingcomparator 7000 at a given cycle. The load current detecting comparator7000 generates a high-voltage level output signal when the voltageVSWH+offset is higher than the ground voltage PGND, and generates alow-voltage level output signal when the voltage VSWH+offset is lowerthan the ground voltage PGND. When output signals from the load currentdetecting comparator 7000 are high-voltage level signals in consecutivefour or more cycles, the counter 7007 outputs a 4 times signal (which isdenoted as 4 times in FIG. 7). When supplied with high-voltage levelsignals from the load current detecting comparator 7000 consecutively inless than four cycles, the counter 7007 outputs a Reset signal (which isdenoted as Reset in FIG. 7).

The RS-type flip-flop 7006 of the four-cycle detecting circuit 7001 hasa set terminal that receives the 4 times signal, and a reset terminalthat receives the Reset signal. When supplied with the 4 times signal,therefore, the RS-type flip-flop 7006 outputs an output signal in itsset state (e.g., high-voltage level state) from the output terminal Q ofthe flip-flop 7006. When supplied with the Reset signal, on the otherhand, the RS-type flip-flop 7006 outputs an output signal in its resetstate (e.g., low-voltage level state) from the output terminal Q of theflip-flop 7006. These output signals from the RS-type flip-flop 7006 areoutput signals from the four-cycle detecting circuit 7001.

The output signal from the four-cycle detecting circuit 7001 is used asa selection signal to the analog switch 7003. The analog switch 7003 hasan n-channel type MOSFET 7008 and a p-channel type MOSFET 7009 havingtheir respective source-drain paths connected in parallel with eachother, and an n-channel type MOSFET 7010 and a p-channel type MOSFET7011 having their respective source-drain paths connected in parallelwith each other. The source-drain paths of the n-channel type MOSFET7008 and p-channel type MOSFET 7009 are connected between the positivevoltage regulator 2000 and the control terminal T14. The source-drainpaths of the n-channel type MOSFET 7010 and p-channel type MOSFET 7011are connected between the negative voltage regulator 2001 and thecontrol terminal T14.

The output signal from the four-cycle detecting circuit 7001 is suppliedto the gate electrode of the p-channel type MOSFET 7011 and to the gateelectrode of the n-channel type MOSFET 7008. The output signal from thefour-cycle detecting circuit 7001 is reversed in phase by the inverter7002 and this phase-reversed output signal is supplied to the gateelectrode of the p-channel type MOSFET 7009 and to the gate electrode ofthe n-channel type MOSFET 7010. Hence according to the output signalfrom the four-cycle detecting circuit 7001, a first analog switchcomposed of the n-channel type MOSFET 7008 and p-channel type MOSFET7009 and a second analog switch composed of the n-channel type MOSFET7010 and p-channel type MOSFET 7011 are switched on and offcomplementally. When the first analog switch is switched on, thepositive voltage Vpos is supplied through the first analog switch to thecontrol terminal T14. When the second analog switch is switched on, thenegative voltage Vneg is supplied through the second analog switch tothe control terminal T14.

The operation of the semiconductor integrated circuit device 4002according to the fourth embodiment will then be described, referring tooperating waveforms shown in FIG. 8. The operation of the semiconductorintegrated circuit device 4002 is summarized as follows. The loadcurrent detecting comparator 7000 detects the load current ISD todetermine whether its value exceeds a given value. Based on a detectionsignal from the load current detecting comparator 7000, the four-cycledetecting circuit 7001 determines whether the value of the load currentISD exceeds the given value in consecutive four or more cycles.According to the result of this determination, the positive voltage Vposor negative voltage Vneg is supplied to the second gate electrode G2 ofthe low-side MOSFET 4006. As a result, the conduction loss is reducedwhen the load is heavy while the switching loss is reduced when the loadis light.

FIGS. 8(A) to 8(D) are waveform charts showing the operation of thesemiconductor integrated circuit device 4002 of FIG. 7. In FIGS. 8(A) to8(D), the horizontal axis represents time. The vertical axes in FIGS.8(A), 8(C), and 8(D) each represent voltage value, and the vertical axisin FIG. 8(B) represents current value.

FIG. 8(A) depicts the waveform of the drive signal GL supplied to thefirst gate electrode G1 of the low-side MOSFET 4006, and FIG. 8(B)depicts the waveform of the source-drain current ISD (load current)flowing through the low-side MOSFET 4006. As describe above, the valueof the source-drain current ISD glows larger as the load becomesheavier.

FIG. 8(C) depicts waveforms supplied to the reverse input terminal (−)and non-reverse input terminal (+) of the comparator 7004, respectively,and FIG. 8(D) depicts an output signal from the four-cycle detectingcircuit 7001 (output signal Q from the RS-type flip-flop 7006).

The drive signal GL output from the drive circuit 4012 goes high involtage level cyclically, thus switching on the low-side MOSFET 4006cyclically. When the low-side MOSFET 4006 is switched on, thesource-drain current ISD is supplied as the load current, through thelow-side MOSFET to the coil element 4008. The flow of the source-draincurrent ISD leads to a drop in the voltage VSWH at the voltage terminalT8 connected to the output terminal T1 (FIG. 40), and leads also to adrop in the voltage VSWH+offset given by adding the offset voltageoffset to the voltage VSWH (VSWH (+offset) in FIG. 8).

In a time zone before time t1, the load is light and the value of theload current (source-drain current ISD) is therefore small. As a result,the voltage VSWH becomes higher than the ground voltage PGND, in whichcase the comparator 7004 supplies a low-voltage level detection signalto the four-cycle detecting circuit 7001. Because the output signal fromthe comparator 7004 is low in voltage level and therefore fails to behigh in voltage level in consecutive four or more cycles, the counter7007 does not generate the 4 time signal. As a result, the output signalQ from the RS-type flip-flop 7006 turns out to be a low-voltage levelsignal. This low-voltage level output signal Q switches the secondanalog switch (MOSFETS 7010 and 7011) on and switches the first analogswitch (MOSFETS 7008 and 7009) off.

Hence, the second gate electrode control signal UL having the negativevoltage Vneg is supplied to the second gate electrode G2 of the low-sideMOSFET 4006. This means that when the load is light, the second gateelectrode control signal UL having the negative voltage Vneg is suppliedto the second gate electrode G2 of the low-side MOSFET 4006 to reducethe switching loss of the low-side MOSFET 4006.

At time t1, when the load increases in size, the value of thesource-drain current ISD flowing through the low-side MOSFET 4006increases. This causes the voltage VSWH at the voltage terminal T8 whenthe low-side MOSFET 4006 is on to drop below the voltage VSWH in thetime zone before time t1. As a result, the voltage VSWH+offset at thereverse input terminal (−) of the comparator 7004 drops below the groundvoltage PGND, in which case the comparator 7004 outputs a high-voltagelevel detection signal. As indicated by FIGS. 8(B) and 8(C), when thesource-drain current ISD remains high in consecutive four cycles, thevoltage VSWH+offset remains lower than the ground voltage PGND inconsecutive four cycles. At the fourth cycle of these consecutivecycles, the comparator 7004 generates the 4 times signal. In response tothis 4 times signal, the flip-flop 7006 is switched to its set state, inwhich case the output signal Q from the flip-flop 7006 goes high involtage level (time t2).

The output signal Q at its high-voltage level switches on the firstanalog switch (MOSFETs 7008 and 7009). As a result, the positive voltageVpos is supplied to the control terminal T14. In other words, the secondgate electrode control signal UL having the positive voltage Vpos issupplied from the terminal T14 to the second gate electrode G2 of thelow-side MOSFET 4006.

In this manner, when the load increases in size to raise the loadcurrent value, that is, the load glows heavier, the positive voltageVpos is supplied to the second gate electrode G2 of the low-side MOSFET4006, which reduces the conduction loss of the low-side MOSFET 4006.

At time t3, when the load reduces in size to become lighter, the loadcurrent value decreases, which leads to a drop in the voltage VSWH atthe voltage terminal T8 when the low-side MOSFET 4006 is on. As aresult, an output signal from the comparator 7004 goes low in voltagelevel, in which case the RS-type flip-flop 7006 is reset, thus producingthe output signal Q that goes low in voltage level at time t4. Hence thesecond gate electrode control signal UL having the negative voltage Vnegis supplied to the second gate electrode G2 again, which reduces theswitching loss of the low-side MOSFET 4006.

As described above, according to the fourth embodiment, a heavy load anda light load are detected. In the case of the heavy load, the secondgate electrode control signal UL having the positive voltage Vpos issupplied to the second gate electrode G2 of the low-side MOSFET 4006 toreduce the conduction loss. In the case of the light load, the secondgate electrode control signal UL having the negative voltage Vneg issupplied to the second gate electrode G2 of the low-side MOSFET 4006 toreduce the switching loss.

According to the fourth embodiment, when the load current is high inconsecutive four or more cycles, the load is determined to be heavy.This method avoids a case where the load is determined to be heavy whenthe load current fluctuates sharply due to noises, etc. Consecutive fourcycles are an example of the number of cycles for load currentevaluation and the number of cycles is not limited to four. Obviously,the load current detecting comparator 7000, the four-cycle detectingcircuit 7001, and the analog switch 7003 may be modified to have variousconfigurations.

According to the above second embodiment, the voltage of the second gateelectrode control signal UL supplied to the second gate electrode G2 ofthe low-side MOSFET 4006 is switched between the positive voltage Vposand the negative voltage Vneg in synchronization with the drive signalGL supplied to the first gate electrode G1 of the low-side MOSFET 4006.In other words, the polarity of the voltage supplied to the second gateelectrode G2 is switched in synchronization with switching on/off of thelow-side MOSFET 4006. According to the third and fourth embodiments, thepolarity of the voltage of the second gate electrode control signal ULsupplied to the second gate electrode G2 of the low-side MOSFET 4006 isswitched according to the load current.

FIGS. 9(A) to 9(F) are explanatory diagrams of the relation between achange in the second gate electrode control signal UL and the heavy loadand light load cases in the third and fourth embodiments.

In FIGS. 9(A) to 9(E), the horizontal axes represent time. FIG. 9(A)depicts the waveform of the output current Iout at the output terminalT1 of the semiconductor integrated circuit device 4002. The outputcurrent Iout includes a current from the high-side MOSFET 4005 and acurrent (source-drain current ISD) from the low-side MOSFET. FIG. 9(B)depicts the waveform of the drive signal GH output from the drivecircuit 4013 to the first gate electrode G1 of the high-side MOSFET4005, and FIG. 9(C) depicts the waveform of the drive signal GL outputfrom the drive circuit 4012 to the first gate electrode G1 of thelow-side MOSFET 4006. FIG. 9(D) depicts the waveform of the outputvoltage (voltage) VSWH at the output terminal T1 (T8). FIG. 9(E) depictsa voltage at the source S of the low-side MOSFET 4006 and the waveformof the second gate electrode control signal UL supplied to the secondgate electrode G2 of the low-side MOSFET 4006. Because the source S ofthe low-side MOSFET 4006 is connected to the ground voltage PGND node,the voltage at the source S matches the ground voltage (GND).

FIG. 9(F) is stacked bar graphs showing losses in the light load caseand losses in the heavy load case.

In FIG. 9, in a time zone before time t1 (left side in FIG. 9), the loadis light and the value of the output current Iout is therefore small.This time zone is thus indicated as “light load” in FIG. 9. In contrast,in a time zone after time t1 (right side in FIG. 9), the load is heavyand the value of the output current Iout is therefore large. This timezone is thus indicated as “heavy load” in FIG. 9. As described abovereferring to FIG. 40, etc., the high-side MOSFET 4005 and low-sideMOSFET are switched on and off alternately by the drive signals GH andGL, in response to which the value of the output voltage (voltage) VSWHat the output terminal T1 (terminal T8) changes.

According to the third and fourth embodiments, the polarity of thevoltage of the second gate electrode control signal UL does not changein synchronization with switching on/off of the low-side MOSFET 4006 butchanges according to the value of the load current (source-drain currentISD). The polarity of the voltage of the second gate electrode controlsignal UL (with respect to a voltage at the source S of the low-sideMOSFET 4006 as a reference voltage) is determined to be negative in thelight load case and to be positive in the heavy load case. As a result,according to the third and fourth embodiments, in a period during whichthe load is light and the low-side MOSFET 4006 is switched on and offseveral times, a negative voltage is supplied constantly to the secondgate electrode G2, as indicated in FIG. 9(E). Likewise, in a periodduring which the load is heavy and the low-side MOSFET 4006 is switchedon and off several times, a positive voltage is supplied constantly tothe second gate electrode G2.

In the light load case, a negative voltage is supplied constantly to thesecond gate electrode G2 of the low-side MOSFET 4006, which reduces thefirst gate-drain capacitance Crss, thereby reduces the switching loss(which is noted as “capacitance reduction/SW loss reduction” in FIG. 9).However, supplying a negative voltage to the second gate electrode G2may increase the on-resistance of the low-side MOSFET 4006 and thereforemay increase the conduction loss (which is noted as “on-resistanceincrease/conduction loss increase” in FIG. 9).

However, as describe above referring to FIG. 38, the ratio of theswitching loss is higher than the ratio of the conduction loss in thelight load case. Based on this fact, as indicated in the time zone onthe left to time t1 in FIG. 9(F), in the light load case, a negativevoltage is supplied to the second gate electrode G2 to reduce theswitching loss so that the overall loss in the light load case can bereduced. In FIG. 9(F), in the time zone in which the load is light(before time t1), a bar graph on the left side to an arrow representsthe breakdown of losses that result when the second gate electrode G2 isconnected to the source S of the low-side MOSFET 4006, while a bar graphon the right side to the arrow represents the breakdown of losses thatresult when, as described in the third and fourth embodiments, anegative voltage is supplied to the second gate electrode G2.

In the heavy load case, a positive voltage is supplied constantly to thesecond gate electrode G2 of the low-side MOSFET 4006, which reduces theon-resistance of the low-side MOSFET 4006, thereby reduces theconduction loss (which is noted as “on-resistance reduction/conductionloss reduction” in FIG. 9). However, supplying a positive voltage to thesecond gate electrode G2 may increase the first gate-drain capacitanceCrss and therefore may increase the switching loss (which is noted as“capacitance increase/SW loss increase” in FIG. 9).

However, as describe above referring to FIG. 38, the ratio of theconduction loss is higher than the ratio of the switching loss in theheavy load case. Based on this fact, as indicated in the time zone onthe right to time t1 in FIG. 9(F), in the heavy load case, a positivevoltage is supplied to the second gate electrode G2 to reduce theconduction loss so that the overall loss in the heavy load case can bereduced. In FIG. 9(F), in the time zone in which the load is heavy(after time t1), a bar graph on the left side to an arrow represents thebreakdown of losses that result when the second gate electrode G2 isconnected to the source S of the low-side MOSFET 4006, while a bar graphon the right side to the arrow represents the breakdown of losses thatresult when, as described in the third and fourth embodiments, apositive voltage is supplied to the second gate electrode G2.

FIG. 10 is a characteristics diagram showing the relation between theoutput current Iout and the efficiency of the semiconductor integratedcircuit device 4002. In FIG. 10, the horizontal axis represents thevalue of the output current Iout and the vertical axis represents theefficiency. In FIG. 10, a broken line represents a characteristics curvethat results when the second gate electrode G2 is connected to thesource S of the low-side MOSFET 4006 (“U-S short”), and a continuousline represents a characteristics curve that results when, as describedin the third and fourth embodiments, a voltage supplied to the secondgate electrode G2 is switched between a positive voltage and a negativevoltage based on the load current. By switching the voltage supplied tothe second gate electrode G2, based on the load current, as described inthe third and fourth embodiments, the efficiency is improved in part ofthe time zone where the load is light as well as part of the time zonewhere the load is heavy. Hence, the overall loss is reduced.

In FIGS. 8 and 9, to make the drawings more understandable, the pulsewidth (high-voltage level period) of the drive signal GL (GH) isdepicted as the same width in both light load case and heavy load case.As a matter of fact, however, the pulse width of the drive signal GL(GH) changes in correspondence to an increase and decrease in the load.Compared to the light load case, a ringing phenomenon of the outputvoltage VSWH becomes more intensive in the heavy load case, as indicatedin FIG. 9.

Losses in the light load case and heavy load case in the first andsecond embodiments will be described.

FIGS. 11(A) to 11(F) are explanatory diagrams of losses in the lightload case and heavy load case that result when a negative voltage issupplied to the second gate electrode G2 of the low-side MOSFET 4006 inthe first embodiment. FIGS. 12(A) to 12(F) are explanatory diagrams oflosses in the light load case and heavy load case that result when apositive voltage is supplied to the second gate electrode G2 of thelow-side MOSFET 4006 in the first embodiment.

FIGS. 11(A) to 11(F) correspond to FIGS. 9(A) to 9(F), respectively, andFIGS. 12(A) to 12(F) also correspond to FIGS. 9(A) to 9(F),respectively. Differences between these figures will mainly bedescribed.

As described above, according to the first embodiment, a negativevoltage or positive voltage is supplied constantly to the second gateelectrode G2 of the low-side MOSFET 4006. As shown in FIGS. 11(E) and12(E), therefore, the second gate electrode control signal UL remains anegative voltage signal or positive voltage signal in both light loadcase and heavy load case.

It is understood from FIG. 11(F) that by supplying a negative voltage tothe second gate electrode G2, the switching loss is reduced in the lightload case, which reduces the overall loss in the light load case. It isunderstood from FIG. 12(F) that by supplying a positive voltage to thesecond gate electrode G2, the conduction loss is reduced, which reducesthe overall loss in the heavy load case.

FIG. 13 is a characteristics diagram showing the output current Ioutfrom the semiconductor integrated circuit device 4002 and the efficiencyof the semiconductor integrated circuit device 4002. Since FIG. 13 issimilar to FIG. 10, differences between FIG. 13 and FIG. 10 will mainlybe described. In FIG. 13, a broken line represents a characteristicscurve that results when the second gate electrode G2 of the low-sideMOSFET 4006 is connected to the source S of the MOSFET 4006. Acontinuous line represents a characteristics curve that results when anegative voltage is supplied to the second gate electrode G2, while asingle-dot broken line represents a characteristics curve that resultswhen a positive voltage is supplied to the second gate electrode G2. Inthis manner, by supplying a negative voltage to the second gateelectrode G2, the efficiency in the light load case is improved. Bysupplying a positive voltage to the second gate electrode G2, on theother hand, the efficiency in the heavy load case is improved.

Connecting the second gate electrode G2 to the source S of the MOSFET isbased on instructions set forth in Patent Document 2.

Losses in the light load case and losses in the heavy load case in thesecond embodiment will then be described. FIGS. 14(A) to 14(F) areexplanatory diagrams of losses in the light load case and heavy loadcase that result when a positive or negative voltage is supplied to thesecond gate electrode G2 of the low-side MOSFET 4006 in the secondembodiment. Since FIGS. 14(A) to 14(F) correspond to FIGS. 9(A) to 9(F),respectively, differences between FIGS. 14(A) to 14(F) and FIGS. 9(A) to9(F) will mainly be described.

As described in the second embodiment, the second gate electrode controlsignal UL changes in synchronization with the drive signal GL suppliedto the first gate electrode G1 of the MOSFET. As shown in FIG. 14(E),the second gate electrode control signal UL becomes a positive voltagesignal when the low-side MOSFET 4006 is switched on by the drive signalGL, and becomes a negative voltage signal when the low-side MOSFET 4006is switched off. When the low-side MOSFET 4006 is switched on,therefore, the on-resistance of the low-side MOSFET is reduced. When thelow-side MOSFET 4006 is switched off, the first gate-drain capacitanceCrss is reduced.

Because the on-resistance can be reduced, the conduction loss thatresults when the low-side MOSFET 4006 is on (which is noted as“on-resistance reduction/conduction loss reduction” in FIG. 14) can bereduced in both light load case and heavy load case. Because the firstgate-drain capacitance Crss can be reduced, the switching loss of thelow-side MOSFET 4006 (which is noted as “capacitance reduction/SW lossreduction” in FIG. 14) can be reduced in both light load case and heavyload case. As a result, as shown in FIG. 14(F), both conduction loss andswitching loss can be reduced in both light load case and heavy loadcase, and therefore the overall loss can be reduced.

FIG. 15 is a characteristic diagram showing the relation betweenefficiency of the semiconductor integrated circuit device and the outputcurrent Iout from the semiconductor integrated circuit device. FIG. 15is similar to FIG. 10 and is different from FIG. 10 in that a continuousline represents a characteristics curve (“positive/negative drive” inFIG. 15) that results when the second gate electrode G2 is drivenaccording to the second embodiment. It is understood from FIG. 10 thataccording to the second embodiment, the efficiency is improved andtherefore the overall loss is reduced in both light load case and heavyload case, compared to a case where the second gate electrode G2 isconnected to the source S (characteristics curve represented by a brokenline).

According to the second embodiment described referring to FIG. 2, theselecting circuit 2002 selects the positive voltage Vpos or negativevoltage Vneg. By improving the withstand voltages of the MOSFETs makingup the selecting circuit 2002, the positive voltage Vpos and negativevoltage Vneg each having a larger absolute value can be supplied to thesecond gate electrode. Using the positive voltage Vpos and negativevoltage Vneg each having a larger absolute value improves the efficiencyindicated in FIG. 15.

Fifth Embodiment

In the first to fourth embodiments relating to the low-side MOSFET 4006,the second gate electrode control signal UL supplied to the second gateelectrode G2 of the low-side MOSFET 4006 has been described. Fifth toeighth embodiments to be described below relate to the high-side MOSFET4005, and a second gate electrode control signal UH supplied to thesecond gate electrode G2 of the high-side MOSFET 4005 will be describedin the fifth to eighth embodiments.

FIG. 16(A) is a circuit diagram showing a configuration of thesemiconductor integrated circuit device 4002 according to the fifthembodiment, and FIG. 16(B) is a waveform chart showing the waveforms ofsignals from the semiconductor integrated circuit device 4002 accordingto the fifth embodiment. FIG. 16(A) depicts the high-side MOSFET 4005and the drive circuit 4011 that outputs the drive signal GH supplied tothe first gate electrode G1 of the high-side MOSFET 4005, out ofcomponents depicted in FIG. 40. Other components in this embodiment arethe same as those of FIG. 40 and are therefore omitted in furtherdescription. According to the fifth embodiment, the driver 4003 isprovided with a control terminal T15. The driver 4003 includes a secondgate electrode control circuit 1600 that supplies the second gateelectrode control signal UH to the second gate electrode G2 of thehigh-side MOSFET 4005 via the control terminal T15.

The second gate electrode control circuit 1600 has a variable voltagesource 1601, and generates the second gate electrode control signal UHhaving a given voltage and supplies the second gate electrode controlsignal UH to the second gate electrode G2 of the high-side MOSFET 4005via the control terminal T15. The given voltage is determined to be, forexample, 2 V, so that the second gate signal UH of 2 V in voltage issupplied from the second gate electrode control circuit 1600 to thesecond gate electrode G2 of the high-side MOSFET 4005 via the controlterminal T15. Needless to say, the first gate electrode G1 of thehigh-side MOSFET 4005 is supplied with the drive signal GH from thedrive circuit 4011. The high-side MOSFET 4005 thus is switched on andoff according to the drive signal GH.

The output terminal T1 of the semiconductor integrated circuit device4002, the voltage terminal T8 of the driver 4003, and the source S ofthe high-side MOSFET 4005 are connected to each other. As describedreferring to FIG. 40, the high-side MOSFET 4005 and low-side MOSFET 4006are switched on and off alternately. As a result, each voltage at theoutput terminal T1, the voltage terminal T8, and the source S of thehigh-side MOSFET 4005 changes time-dependently. In other words,switching on/off of the high-side MOSFET 4005 and low-side MOSFET 4006causes the voltage at the source S of the high-side MOSFET 4005 tochange.

Complemental switching on/off of the high-side MOSFET 4005 and low-sideMOSFET 4006 causes the voltage VSWH at the voltage terminal T8 to changeroughly between the ground voltage (GND=0 V) and the voltage VIN. Thevoltage amplitude representing this voltage change is determined to be,for example, 12 V and the voltage of the second control signal UH isdetermined to be 2 V, as described above. In this case, at a point oftime right before switching on of the high-side MOSFET 4005, the voltageat the voltage terminal T8 is almost equal to the ground voltage becausethe low-side MOSFET 4006 is on, so that a positive voltage of 2 Vpositive in polarity relative to the voltage at the source S (voltageterminal T8) is supplied to the second gate electrode G2 of thehigh-side MOSFET 4005. At a point of time right before switching off ofthe high-side MOSFET 4005, the voltage at the source S (voltage terminalT8) of the MOSFET 4005 is 12 V because it is on, so that a negativevoltage of −10 V negative in polarity relative to the voltage at thesource S of the MOSFET 4005 is supplied to the second gate electrode G2of the high-side MOSFET 4005.

FIG. 16(B) depicts the waveform of the voltage VSWH (source voltage(VSWH)) at the source S (voltage terminal T8) of the high-side MOSFET4005 and the waveform of the second control signal UH. In FIG. 16(B),the horizontal axis represents time and the vertical axis representsvoltage value. In FIG. 16(B), a period in which the high-side MOSFET4005 is switched on is indicated as a period (a), while a period inwhich the high-side MOSFET 4005 is switched off is indicated as a period(b). FIG. 16(B) also depicts a period in which the voltage VSWH at thesource S (source voltage) drops below the ground voltage (0 V). Thisperiod indicates a voltage change caused by a counter electromotiveforce generated by the coil element 4008 (FIG. 40).

According to the fifth embodiment, in the period in which the high-sideMOSFET 4005 is switched off (period (b)), a negative voltage negative inpolarity relative to the source voltage (VSWH) at the source S of theMOSFET 4005 is supplied to the second gate electrode G2 of the high-sideMOSFET 4005, in which case the first gate-drain capacitance Crss of thehigh-side MOSFET 4005 is reduced. As a result, the switching loss isreduced.

FIGS. 17(A) to 17(E) are explanatory diagrams of losses in the lightload case and heavy load case that result when the second gate controlsignal UH having the given voltage is supplied from the second gateelectrode control circuit 1600 to the second gate electrode G2 of thehigh-side MOSFET 4005 in the fifth embodiment. FIGS. 17(A) to 17(C) andFIG. 17(E) correspond to FIGS. 9(A) to 9(C) and FIG. 9(F), respectively.FIG. 17(D) is similar to FIG. 9(D) and to FIG. 9(E), and depicts thewaveform of the voltage (source voltage) VSWH at the source S (voltageterminal T8) of the high-side MOSFET 4005 and the waveform of the secondgate control signal UH. Differences between FIG. 17 and FIG. 9 willmainly be described.

As described referring to FIGS. 16(A) and 16(B), according to the fifthembodiment, the positive given voltage is generated by the second gateelectrode control circuit 1600 and is supplied constantly to the secondgate electrode G2 of the high-side MOSFET 4005. As a result, when thehigh-side MOSFET 4005 is switched off, a negative voltage negative inpolarity relative to a voltage at the source of the high-side MOSFET4005 is supplied to the second gate electrode G2. Because this negativevoltage is supplied constantly, the high-side MOSFET 4005 can reduce itswitching loss when shifting from its on-state to off-state in both lightload case and heavy load case (which are noted as “(Turn Off)capacitance reduction/SW loss reduction”).

As indicated in FIG. 17(E), a reduction in the switching loss leads to areduction in the overall loss in the light load case. This is becausethat, as described referring to FIG. 38, the ratio of the switching lossis high in the light load case. However, when the high-side MOSFET 4005is switched on, the first gate-drain capacitance Crss may increases. Inaddition, in a period during which the high-side MOSFET 4005 remains on,a voltage supplied to the second gate electrode G2 is negative inpolarity relative to a voltage at the source S, which can be understoodby observing FIG. 17(D), and consequently the on-resistance mayincrease.

FIG. 18 is a characteristic diagram showing the relation between theefficiency of the semiconductor integrated circuit device 4002 of thefifth embodiment and the output current Iout from the semiconductorintegrated circuit device. FIG. 18 is similar to FIG. 10 and isdifferent from FIG. 10 in that the relation between the output currentIout from the semiconductor integrated circuit device 4002 of the fifthembodiment and the efficiency of the semiconductor integrated circuitdevice 4002 is indicated by a continuous line (“constant voltage”). FIG.18 demonstrates that the efficiency is improved and therefore theoverall loss is reduced in the light load case.

Sixth Embodiment

FIG. 19(A) is a circuit diagram showing a configuration of thesemiconductor integrated circuit device 4002 according to a sixthembodiment, and FIG. 19(B) is a waveform chart showing the waveforms ofsignals from the semiconductor integrated circuit device 4002 of FIG.19(A). Since the configuration shown in FIG. 19(A) is similar to theconfiguration shown in FIG. 16(A) described in the fifth embodiment,differences between both configurations will mainly be described.

The configuration of the second gate electrode control circuit 1600shown in FIG. 19(A) is different from the configuration of the secondgate electrode control circuit 1600 shown FIG. 16(A). In FIG. 19(A), thesecond gate electrode control circuit 1600 has resistance elements 1900and 1901 connected in series between the voltage terminal T8 and theground voltage CGND node. The voltage (source voltage) VSWH at thesource of the high-side MOSFET 4005 is divided through the resistanceelements 1900 and 1901. A voltage resulting from the voltage division isextracted from a connection node between the resistance element 1900 andthe resistance element 1901 and is supplied as the second gate controlsignal UH, from the second gate electrode control circuit 1600 to thecontrol terminal T15, which is connected to the second gate electrode G2of the high-side MOSFET 4005. As a result, a voltage corresponding tothe voltage VSWH at the source S of the high-side MOSFET 4005 issupplied as the second gate control signal UH, to the second gateelectrode G2 of the high-side MOSFET 4005.

FIG. 19(B) depicts the waveform of the source voltage VSWH at the sourceS of the high-side MOSFET 4005 (voltage at the voltage terminal T8) ofFIG. 19(A) and the waveform of the second gate control signal UHgenerated by the voltage division. In FIG. 19(B), a period (a)represents a period in which the high-side MOSFET 4005 shifts from itsoff-state to on-state, and a period (b) represents a period in which thehigh-side MOSFET 4005 shifts from its on-state to off-state. Because thesecond gate control signal UH is generated by dividing the voltage atthe source of the high-side MOSFET 4005, the voltage of the second gatecontrol signal UH changes by following a change in the voltage VSWH atthe source.

When the high-side MOSFET 4005 in its off-state is switched on (period(a)), therefore, a voltage rising in the same manner as the voltage VSWHat the source of the MOSFET 4005 is supplied to the second gateelectrode G2 of the high-side MOSFET 4005. Hence, when the high-sideMOSFET 4005 shifts from its off-state to on-state, a voltage differencebetween the voltage at the second gate electrode G2 and the voltage atthe source of the high-side MOSFET 4005 is reduced, which suppresses anincrease in the first gate-drain capacitance Crss. When the high-sideMOSFET 4005 in its on-state is switched off (period (b)), because thevoltage of the second gate control signal UH is the divided voltagegenerated out of the source voltage VSWH of the high-side MOSFET 4005, avoltage supplied to the second gate electrode G2 is lower than thesource voltage VSWH and is negative in polarity relative to the sourcevoltage VSWH. Hence, when the high-side MOSFET 4005 shifts from itson-state to off-state, the first gate-drain capacitance Crss is reducedand therefore the switching loss is reduced in the same manner as in thefifth embodiment.

According to the sixth embodiment, an increase in the first gateelectrode-drain capacitance Crss can be suppressed when the high-sideMOSFET 4005 in its off-state is switched on, and the first gateelectrode-drain capacitance Crss can be reduced when the high-sideMOSFET 4005 in its on-state is switched off. Hence, in the same manneras in the fifth embodiment, the switching loss of the high-side MOSFETcan be reduced.

FIGS. 20(A) to 20(E) are explanatory diagrams of losses in the lightload case and heavy load case of the semiconductor integrated circuitdevice 4002 of the sixth embodiment. FIGS. 20(A) to 20(E) correspond toFIGS. 17(A) to 17(E), respectively. Differences between FIGS. 20(A) to20(E) and FIGS. 17(A) to 17(E) will mainly be described.

A waveform shown in FIG. 20(D) is different from the waveform shown inFIG. 17(D). According to the sixth embodiment, the voltage of the secondgate control signal UH supplied to the second gate electrode G2 of thehigh-side MOSFET 4005 changes by following a change in the voltage VSWHat the source of the high-side MOSFET 4005. The voltage amplitude of thesecond gate control signal UH is smaller than the amplitude of thevoltage (source voltage VSWH) at the source of the high-side MOSFET4005.

As a result, when the high-side MOSFET 4005 shifts from its on-state tooff-state, a negative voltage negative in polarity relative to thesource voltage is supplied to the second gate electrode G2 to reduce theswitching loss (which is noted as (Turn Off) capacitance reduction/SWloss reduction in FIG. 20(D)). Because the voltage supplied to thesecond gate electrode G2 follows the source voltage VSWH, when thehigh-side MOSFET 4005 shifts from its on-state to off-state, a voltagedifference between the voltage at the second gate electrode G2 and thesource voltage VSWH is reduced, which suppresses an increase in thefirst gate-drain capacitance Crss (which is noted as “(Turn On) nocapacitance change”). These actions take place in both light load caseand heavy load case. Hence, the switching loss of the high-side MOSFET4005 is reduced in both light load case and heavy load case (see FIG.20(E)).

FIG. 21 is a characteristic diagram showing the relation between theefficiency of the semiconductor integrated circuit device 4002 of thesixth embodiment and the output current Iout from the semiconductorintegrated circuit device. FIG. 21 is similar to FIG. 18 and isdifferent from FIG. 18 in that the relation between the output currentIout from the semiconductor integrated circuit device 4002 of the fifthembodiment and the efficiency of the semiconductor integrated circuitdevice 4002 is indicated by a single-dot broken line (“constantvoltage”) and that the relation between the output current Iout from thesemiconductor integrated circuit device 4002 of the sixth embodiment andthe efficiency of the semiconductor integrated circuit device 4002 isindicated by a continuous line (VSWH voltage division). It is understoodfrom FIG. 21 that when the output current Iout is small, that is, whenthe load is light, the efficiency is improved and therefore overall lossis reduced.

Seventh Embodiment

FIG. 22(A) is a circuit diagram showing a configuration of thesemiconductor integrated circuit device 4002 according to a seventhembodiment, and FIG. 22(B) is a waveform chart showing the waveforms ofsignals from the semiconductor integrated circuit device 4002 of FIG.22(A). Since the configuration shown in FIG. 22(A) is similar to theconfiguration shown in FIG. 19(A) described in the sixth embodiment,differences between both configurations will mainly be described.

The configuration of the second gate electrode control circuit 1600shown in FIG. 22(A) is different from the configuration of the secondgate electrode control circuit 1600 shown in FIG. 19(A). The second gateelectrode control circuit 1600 of FIG. 19(A) generates a divided voltagethrough the resistance elements 1900 and 1901 and supplies the generateddivided voltage as the second gate control signal UH. The second gateelectrode control circuit 1600 of the seventh embodiment has aresistance element 2200 connected between the voltage terminal T8 andthe control terminal T15, a variable resistance element 2201 connectedbetween the control terminal T18 and the ground voltage CGND node, ahigh-side voltage control circuit 2203 for changing the resistance valueof the variable resistance element 2201, and a load current detectingcircuit 2202.

The load current detecting circuit 2202 detects the output current Ioutflowing through the output terminal T1 of the semiconductor integratedcircuit device 4002 to determine whether the value of the output currentIout exceeds a given current value and outputs a detection signalindicating whether the value of the output current Iout exceeds thegiven current value to the high-side voltage control circuit 2203.According to the supplied detection signal, the high-side voltagecontrol circuit 2203 changes the resistance value of the variableresistance element 2201. The resistance element 2200 and the variableresistance element 2201 are connected in series between the outputterminal T8 and the ground voltage CGND node, and a connection nodebetween both resistance elements is connected to the control terminalT15. Hence the voltage (source voltage) VSWH at the source S of thehigh-side MOSFET 4005 is divided through the resistance element 2200 andthe variable resistance element 2201, and the second gate control signalUH having a divided voltage generated by the voltage division issupplied to the second gate electrode G2 of the high-side MOSFET 4005via the control terminal T15. The resistance value of the variableresistance element 2201 is changed by the high-side voltage controlcircuit 2203 according to the detection signal from the load currentdetecting circuit 2202. In other words, the value of the voltage signalsupplied to the second gate electrode G2 of the high-side MOSFET 4005changes according to the value of the load current.

According to the seventh embodiment, when the value of the outputcurrent Iout (load current) exceeds the given current value, that is,when the value of an output signal from the load current detectingcircuit 2202 exceeds a given value, the high-side voltage controlcircuit 2203 increases the resistance value of the variable resistanceelement 2201. To put it another way, the resistance value of thevariable resistance element 2201 is determined to be a first resistancevalue when the value of the output current Iout is equal to or smallerthan the given current value, and is determined to be a secondresistance value larger than the first resistance value when the valueof the output current Iout exceeds the given current value. Hence, whenthe value of the output current Iout exceeds the given current value, avoltage higher than a voltage for the case of the output current Iout ofthe given current value or smaller current value is supplied as thesecond gate control signal UH, to the second gate electrode G2 of thehigh-side MOSFET 4005. Specifically, when the value of the outputcurrent Iout exceeds the given current value, a voltage close to thesource voltage VSWH of the high-side MOSFET 4005 is supplied to thesecond gate electrode G2 of the high-side MOSFET 4005.

The given current value is the load current value with respect to whichthe light load case and the heavy load case are distinguished from eachother. For example, in the case of FIG. 38, the current value i2 isequivalent to the given current value. Under this setting, when thehigh-side MOSFET 4005 is on in the heavy load case, the second gatecontrol signal UH having a voltage close to the source voltage VSWH issupplied to the second gate electrode G2 of the high-side MOSFET 4005.Hence, an increase in the on-resistance is suppressed when the high-sideMOSFET 4005 is on.

FIG. 22(B) depicts the waveform of the source voltage VSWH at the sourceS of the high-side MOSFET 4005 (voltage at the voltage terminal T8) ofFIG. 22(A) and the waveform of the second gate control signal UHgenerated by the voltage division through the variable resistanceelement 2201 and resistance element (stationary resistance element)2200. Since the waveforms shown in FIG. 22(B) are similar to thewaveforms shown in FIG. 19(B), differences between the waveforms of FIG.22(B) and the waveforms of the FIG. 19(B) will mainly be described. InFIG. 22(B), a period (a) represents a period in which the high-sideMOSFET 4005 shifts from its off-state to on-state, and a period (b)represents a period in which the high-side MOSFET 4005 shifts from itson-state to off-state. The action made in these periods (a) and (b) isdepicted as the same action in FIG. 22(B) and FIG. 19(B), which meansthat the voltage of the second gate control signal UH changes byfollowing a change in the voltage VSWH at the source.

Hence, when the high-side MOSFET 4005 shifts from its off-state toon-state (period (a)), a voltage difference between the voltage at thesecond gate electrode G2 and the voltage at the source of the high-sideMOSFET 4005 is reduced, which suppresses an increase in the firstgate-drain capacitance Crss. When the high-side MOSFET 4005 shifts fromits on-state to off-state (period (b)), because the voltage of thesecond gate control signal UH is the divided voltage generated out ofthe source voltage VSWH of the high-side MOSFET 4005, a voltage suppliedto the second gate electrode G2 is lower than the source voltage VSWHand is negative in polarity relative to the source voltage VSWH. Hence,when the high-side MOSFET 4005 shifts from its on-state to off-state,the first gate-drain capacitance Crss is reduced and therefore theswitching loss is reduced.

According to the seventh embodiment, when the value of the load currentIout exceeds the given value, the high-side voltage control circuit 2203increases the resistance value of the variable resistance element 2201.As a result, when the high-side MOSFET 4005 is on, a voltage supplied tothe second gate electrode G2 approaches the source voltage VSWH of thehigh-side MOSFET 4005. This suppresses an increase in the on-resistanceof the high-side MOSFET 4005, thereby suppresses an increase in theconduction loss. Hence, compared to the fifth and sixth embodiments, theconduction loss of the high-side MOSFET is reduced further. In thismanner, according to the seventh embodiment, an increase in theconduction loss can be suppressed in the heavy load case where the ratioof the conduction loss is high.

FIGS. 23(A) to 23(E) are explanatory diagrams of losses in the lightload case and heavy load case of the semiconductor integrated circuitdevice 4002 of the seventh embodiment. FIGS. 23(A) to 23(E) correspondto FIGS. 20(A) to 20(E), respectively. Differences between FIGS. 23(A)to 23(E) and FIGS. 20(A) to 20(E) will mainly be described.

A waveform shown in FIG. 23(D) is different from the waveform shown inFIG. 20(D) in the heavy load case. According to the seventh embodiment,based on an incoming detection signal from the load current detectingcircuit 2202, the high-side voltage control circuit 2203 recognizes thatthe value of the load current Iout has exceeded the given value, thusincreasing the resistance value of the variable resistance element 2201.When the value of the load current Iout is equal to or smaller than thegiven value, i.e., in the light load case, the high-side voltage controlcircuit 2203 does not increase the resistance value of the variableresistance element 2201. As a result, in the light load case, theswitching loss of the high-side MOSFET 4005 is reduced in the samemanner as in the sixth embodiment (see bar graphs in the light load casein FIG. 23(E)).

In the heavy load case, the voltage of the second gate control signal UHsupplied to the second gate electrode G2 of the high-side MOSFET changesin the same manner as the source voltage VSWH of the high-side MOSFET4005 changes, thus turning into a voltage close to the source voltageVSWH. In other words, in the heavy load case, the potential state of thesecond gate electrode G2 of the high-side MOSFET 4005 comes closer to astate of being shorted to the source S of the high-side MOSFET 4005(which is noted as “close to US short (U-S short)” in FIG. 23(D)). As aresult, in the heavy load case, an increase in the on-resistance of thehigh-side MOSFET 4005 is suppressed and therefore an increase in theconduction loss is suppressed. Hence, according to the seventhembodiment, the switching loss can be reduced in the light load casewhere its ratio is high and an increase in the conduction loss issuppressed in the heavy load case where its ratio is high.

FIG. 24 is a characteristic diagram showing the relation between theefficiency of the semiconductor integrated circuit device 4002 of theseventh embodiment and the output current Iout from the semiconductorintegrated circuit device. FIG. 24 is similar to FIG. 21 and isdifferent from FIG. 21 in that the relation between the output currentIout from the semiconductor integrated circuit device 4002 of the sixthembodiment and the efficiency of the semiconductor integrated circuitdevice 4002 is indicated by a single-dot broken line (“VSWH voltagedivision”) and that the relation between the output current Iout fromthe semiconductor integrated circuit device 4002 of the seventhembodiment and the efficiency of the semiconductor integrated circuitdevice 4002 is indicated by a continuous line (“voltage division+voltagedivision ratio control”). It is understood from FIG. 24 that when thevalue of the output current Iout is large, i.e., in the heavy load case,the efficiency is improved and therefore the overall loss is reducedfurther, compared to the sixth embodiment.

In the case of FIG. 22, the resistance element connected between thecontrol terminal T15 and the ground voltage CGND node functions as thevariable resistance element. However, the resistance element connectedbetween the control terminal T15 and the ground voltage CGND node may beprovided as a stationary resistance element and the resistance elementconnected between the output terminal T8 and the control terminal T15may be provided as a variable resistance element whose resistance valueis controlled by the high-side voltage control circuit 2203. In such acase, the resistance value of the variable resistance element iscontrolled to be small in the heavy load case. In another configuration,both of the resistance elements 2200 and 2201 may be provided asvariable resistance elements whose respective resistance values arecontrolled by the high-side voltage control circuit 2203.

The resistance value of the variable resistance element may not beswitched between two resistance values, i.e., first resistance value andsecond resistance value, but may be switched between three or moreresistance values. The resistance value of the variable resistanceelement may also be changed in such a way that the load currentdetecting circuit 2202 outputs a detection signal whose value changescontinuously according to the load current so that the high-side voltagecontrol circuit 2203 changes the resistance value of the variableresistance element continuously according to this detection signal.

Eighth Embodiment

FIG. 25 is a waveform diagram showing the waveform of the second gatecontrol signal UH generated by the second gate electrode control circuit1600 included in the semiconductor integrated circuit device 4002according to an eighth embodiment. The second gate control signal UHgenerated by the second gate electrode control circuit 1600 of theeighth embodiment is supplied to the second gate electrode G2 of thehigh-side MOSFET 4005, for example, via the control terminal T15 of FIG.22.

In FIG. 25, the horizontal axis represents time and the vertical axisrepresents voltage. When the high-side MOSFET 4005 is kept switched offby the drive signal GH supplied to the first gate electrode G1 of thehigh-side MOSFET 4005, the second gate electrode control circuit 1600 ofthe eighth embodiment supplies the second gate control signal UH havinga negative voltage V1 negative in polarity relative to the sourcevoltage VSWH at the source S of the high-side MOSFET 4005 (voltage atthe output terminal T8), to the second gate electrode G2 of thehigh-side MOSFET 4005.

At time t1 right before a point of time at which the high-side MOSFET4005 is shifted from its off-state to on-state by the drive signal GH,the second gate electrode control circuit 1600 changes the voltage ofthe second gate control signal UH, from the voltage V1 to a voltage V2.This voltage V2 is determined to be higher than the source voltage VSWHof the high-side MOSFET 4005 in its on-state.

Subsequently, at time t2 right before a point of time at which thehigh-side MOSFET 4005 is shifted from its on-state to off-state by thedrive signal GH, the second gate electrode control circuit 1600 changesthe voltage of the second gate control signal UH, from the voltage V2 toa voltage V3. This voltage V3 is determined to be negative in polarityrelative to the source voltage VSWH of the high-side MOSFET 4005. Thesecond gate electrode control circuit 1600 changes the voltage of thesecond gate control signal UH to the voltage V3 and then further changesthe voltage V3 to the voltage V1 (the voltage shift of the second gatecontrol signal UH is not limited to this).

By changing the voltage of the second gate control signal UHsequentially in this manner, when the high-side MOSFET 4005 shifts fromits off-state to on-state (time t1), a negative voltage negative inpolarity relative to the source voltage VSWH is supplied to the secondgate electrode G2. In the same manner, when the high-side MOSFET 4005shifts from its on-state to off-state (time t2), a negative voltagenegative in polarity relative to the source voltage VSWH of thehigh-side MOSFET 4005 is supplied to the second gate electrode G2.Hence, when the high-side MOSFET 4005 shifts from its off-state toon-state or from its off-state to on-state, the first gate-draincapacitance Crss of the high-side MOSFET 4005 is reduced and thereforethe switching loss is reduced.

In a period during which the high-side MOSFET 4005 remains on (periodbetween time t1 and time t2), a positive voltage positive in polarityrelative to the source voltage VSWH is supplied to the second gateelectrode G2, which reduces the on-resistance of the high-side MOSFET4005, thereby reduces the conduction loss. Hence, both switching lossand conduction loss can be reduced.

The second gate electrode control circuit 1600 that generates the secondgate control signal UH changing in voltage in the manner as shown inFIG. 25 can be realized using, for example, a negative voltagegenerating circuit that generates the voltage V1, a positive voltagegenerating circuit that generates the voltage V2, and a logical circuitthat receives the control signal f from the driver 4004 describedreferring FIG. 40. For example, based on the control signal f, a shiftin the voltage of the drive signal GH supplied to the first gate G1 ofthe high-side MOSFET 4005 to a high-voltage level is grasped before thevoltage shift occurs, and the voltage of the second gate control signalUH is changed to the voltage V2 in advance. In the same manner, based onthe control signal f, a shift in the voltage of the drive signal GH to alow-voltage level is grasped before the voltage shift occurs, and thevoltage of the second gate control signal UH is changed from the voltageV2 to the voltage V3 in advance. The voltage V3 can be generated fromthe voltage V2. The voltage V3 may be determined by measurement or basedon a measured voltage at the output terminal T8.

FIG. 26 is a characteristic diagram showing the relation between theefficiency of the semiconductor integrated circuit device 4002 of theeighth embodiment and the output current Iout from the semiconductorintegrated circuit device. FIG. 26 is similar to FIG. 24 described aboveand is different from FIG. 24 in that the relation between the outputcurrent Iout from the semiconductor integrated circuit device 4002 ofthe seventh embodiment and the efficiency of the semiconductorintegrated circuit device 4002 is indicated by a single-dot broken line(“voltage division+voltage division ratio control”) and that therelation between the output current Iout from the semiconductorintegrated circuit device 4002 of the eighth embodiment and theefficiency of the semiconductor integrated circuit device 4002 isindicated by a continuous line (“eighth embodiment”). According to theeighth embodiment, both switching loss and conduction loss are reduced.It is understood from FIG. 26 that in both cases of the output currentIout being large and small, i.e., in both heavy load case and light loadcase, the efficiency is improved and therefore the overall loss isreduced.

Ninth Embodiment

FIG. 27 is a block diagram showing a configuration of the semiconductorintegrated circuit device 4002 according to a ninth embodiment. Thesemiconductor integrated circuit device 4002 of FIG. 27 is similar tothe semiconductor integrated circuit device 4002 described referring toFIG. 40. In both FIGS. 27 and 40, the same constituent elements aredenoted by the same reference numerals. Only the constituent elementsdifferent between FIG. 27 and FIG. 40 will mainly be described.

Being different from the driver 4003 of FIG. 40, the driver 4003 of theninth embodiment includes a load current detecting circuit 2700, asecond gate electrode control circuit 2701, the control terminal T14,and the control terminal T15.

The load current detecting circuit 2700 is connected to the voltageterminal T2 of the semiconductor integrated circuit device 4002 via thevoltage terminal T10 of the driver 4003 and is connected to the outputterminal T1 of the semiconductor integrated circuit device 4002 via thevoltage terminal T8 of the driver 4003. The load current detectingcircuit 2700 is equivalent to the load current detecting circuit(including the load current detecting comparator 7000 (FIG. 7))described in the above multiple embodiments. For example, the loadcurrent detecting circuit 2700 is equivalent to the load currentdetecting circuit 5000 (FIG. 5) or the load current detecting circuit2202 (FIG. 22) described in the third or seventh embodiment. The loadcurrent detecting circuit 2700 detects the output current Iout flowingthrough the output terminal T1 of the semiconductor integrated circuitdevice 4002, as a load current, determines whether the value of theoutput current Iout has exceeded a given current value (e.g., thecurrent i2 of FIG. 38), and supplies a detection signal to the secondgate electrode control circuit 2701.

According to the detection signal from the load current detectingcircuit 2700, the second gate electrode control circuit 2701 generatesthe second gate control signal UH and the second gate control signal UL.The generated second gate control signal UH is used to control thesecond gate electrode G2 of the high-side MOSFET 4005, while the secondgate control signal UL is used to control the second gate electrode G2of the low-side MOSFET 4006. Therefore, the second gate control signalUH is supplied to the second gate electrode G2 of the high-side MOSFET4005 via the control terminal T15, and the second gate control signal ULis supplied to the second gate electrode G2 of the low-side MOSFET 4006via the control terminal T14.

When the detection signal from the load current detecting circuit 2700indicates that the value of the output current Iout flowing through theoutput terminal T1 exceeds the given current value, the second gateelectrode control circuit 2701 generates the second gate control signalUH having a positive voltage positive in polarity relative to thevoltage VSWH at the source S of the high-side MOSFET 4005. In this case,the second gate electrode control circuit 2701 generates the second gatecontrol signal UL having a positive voltage positive in polarityrelative to the voltage PGND at the source S of the high-side MOSFET4006.

When the detection signal indicating that the value of the outputcurrent Iout is equal to or smaller than the given current value issupplied to the second gate electrode control circuit 2701, the secondgate electrode control circuit 2701 generates the second gate controlsignal UH having a negative voltage negative in polarity relative to thevoltage VSWH at the source S of the high-side MOSFET 4005 and the secondgate control signal UL having a negative voltage negative in polarityrelative to the voltage PGND at the source S of the high-side MOSFET4006.

When the load, such as a CPU, becomes heavier, the load current (outputcurrent) Iout increases. In this embodiment, the value of the current i2is set as the load current value with respect to which the light loadcase and the heavy load case are distinguished from each other. Hence,in the heavy load case, a (positive) voltage higher than the voltage atthe source S of the high-side MOSFET 4005 is supplied constantly to thesecond gate electrode G2 of the high-side MOSFET 4005. In the samemanner, a (positive) voltage higher than the voltage at the source S ofthe low-side MOSFET 4006 is supplied constantly to the second gateelectrode G2 of the low-side MOSFET 4006. Because the voltage suppliedto the second gate electrode G2 is positive in polarity relative to thevoltage at the source, the on-resistance of the high-side MOSFET 4005and low-side MOSFET 4006 in their on-state is reduced. As a result,respective conduction losses of the high-side MOSFET 4005 and low-sideMOSFET 4006 in the heavy load case are reduced.

When the value of the load current Iout is equal to or smaller than thegiven current value (i2) (in the light load case), a negative voltagenegative in polarity relative to a voltage at the source S of thehigh-side MOSFET 4005 is supplied constantly to the second gateelectrode G2 of the high-side MOSFET 4005 and a negative voltagenegative in polarity relative to a voltage at the source S of thelow-side MOSFET 4006 is supplied constantly to the second gate electrodeG2 of the low-side MOSFET 4006. As a result, in the light load case,respective first gate electrode-drain capacitances Crss of the high-sideMOSFET 4005 and low-side MOSFET 4006 are reduced and therefore theswitching loss is reduced.

The second gate electrode control circuit 2701 is composed of a positivevoltage regulator, a negative voltage regulator, and four switches (theconfiguration of the second gate electrode control circuit 2701 is,however, not limited to this). The positive regulator generates, forexample, the positive voltage Vpos described referring to FIG. 5 and thepositive voltage V2 described referring to FIG. 25. The negativeregulator generates the negative voltage Vneg described referring toFIG. 5 and the negative voltage V1 described referring to FIG. 25. Thefour switches are two pairs of switches. In the same manner as in thecase of the switches 5004 and 5005 of FIG. 5, one ends of one pair ofswitches are supplied with the positive voltage Vpos and the negativevoltage Vneg, respectively, and the second gate control signal UL isoutput from the other ends of one pair of switches. Likewise, one endsof the other pair of switches are supplied with the positive voltage V2and the negative voltage V1, respectively, and the second gate controlsignal UH is output from the other ends of the other pair of switches.Hence, the two pairs of switches are controlled based on whether thedetection signal from the load current detecting circuit 2700 exceedsthe given value in order to select a voltage supplied to respectivesecond gate electrodes G2 of the high-side MOSFET 4005 and low-sideMOSFET 4006.

Specifically, the switches supplied with the positive voltages Vpos andV2 are switched on in the heavy load case, and the switches suppliedwith the negative voltages Vneg and V1 are switched on in the light loadcase. Needless to say, when the switches supplied with the positivevoltages Vpos and V2 (negative voltages Vneg and V1) are switched on,the switches supplied with the negative voltages Vneg and V1 (positivevoltages Vpos and V2) are switched off.

According to the ninth embodiment, the conduction loss and switchingloss are reduced according to the condition of the load. In the heavyload case, the conduction loss whose ratio is high in the heavy loadcase can be reduced at both high-side MOSFET 4005 and low-side MOSFET4006, and in the light load case, the switching loss whose ratio is highin the light load case can be reduced at both high-side MOSFET 4005 andlow-side MOSFET 4006. To put it another way, proper loss reductionaccording to the condition of the load at the time of loss reduction iscarried out so that the overall loss can be reduced regardless of achange in the load condition.

First Modification Example

According to the description of FIG. 27, in the heavy load case, avoltage higher than a voltage at the source S of the high-side MOSFET4005 is supplied constantly to the second gate electrode G2 of thehigh-side MOSFET 4005. However, the configuration of the seventhembodiment may be applied to the high-side MOSFET 4005 of FIG. 27, inwhich case the configuration of the third embodiment is applied to thelow-side MOSFET 4006 of FIG. 27.

In such a case, the load current detecting circuit 2700 of FIG. 27 has,for example, the load current detecting circuit 5000 of FIG. 5 and theload current detecting circuit 2202 of FIG. 22. The second gateelectrode control circuit 2701 of FIG. 27 has the second gate electrodedrive circuit 5001, positive voltage regulator 5002, negative voltageregulator 5003, and switches 5004 and 5005 of FIG. 5 and the high-sidevoltage control circuit 2203, resistance element 2200, and variableresistance element 2201 of FIG. 22(A).

According to the first modification example, in the heavy load case, avoltage at the second gate electrode G2 of the high-side MOSFET 4005changes by following a change in a voltage at the source S of thehigh-side MOSFET. For this reason, a voltage regulator that generatesthe positive voltage V2 and negative voltage V1 supplied to the secondgate electrode G2 of the high-side MOSFET 4005 can be dispensed with.

Second Modification Example

FIG. 28 is a block diagram showing a configuration of the semiconductorintegrated circuit device 4002 according to a modification example ofthe ninth embodiment. The configuration of FIG. 28 is similar to theconfiguration of FIG. 27. Differences between both configurations,therefore, will mainly be described.

In FIG. 28, 2802 denotes a MOSFET having no second gate electrode G2.The MOSFET 2802 of such a structure is known as, for example, atrench-type MOSFET, which is, for example, a MOSFET constructed byeliminating the insulating layer and the metal layer 3708 equivalent tothe second gate electrode G2 from the n-type semiconductor layer 3704 ofFIG. 37(B) (hereinafter “single-gate electrode MOSFET”). In FIG. 28,2800 denotes a load current detecting circuit and 2801 denotes a secondgate electrode control circuit.

The load current detecting circuit 2800 detects the output current (loadcurrent) Iout flowing through the output terminal T1 and supplies adetection signal indicating whether the value of the output current Ioutexceeds a given current value, to the second gate electrode controlcircuit 2801. The second gate electrode control circuit 2801 generatesthe second gate control signal UL according to the detection signal andsupplies the second gate control signal UL to the second gate electrodeG2 of the low-side MOSFET 4006 via the control terminal T14. In thesemiconductor integrated circuit device 4002 of FIG. 28, because thehigh-side MOSFET is the single-gate electrode MOSFET 2802, the secondgate control signal UH for the high-side MOSFET is not generated.

The configuration described in the third or fourth embodiment is appliedto the load current detecting circuit 2800 and the second gate electrodecontrol circuit 2801.

When the configuration of the third embodiment is applied, the secondgate electrode drive control circuit 5001, positive voltage regulator5002, negative voltage regulator 5003, and switches 5004 and 5005 ofFIG. 5 are collectively regarded as the second gate electrode controlcircuit 2801 of FIG. 28. The load current detecting circuit 5000 of FIG.5 is regarded as the load current detecting circuit 2800 of FIG. 28.

When the configuration of the fourth embodiment is applied, the loadcurrent detecting comparator 7000 of FIG. 7 is regarded as the loadcurrent detecting circuit 2800 of FIG. 28. The four-cycle detectingcircuit 7001, analog switch 7003, inverter 7002, positive voltageregulator 2000, and negative voltage regulator 2001 of FIG. 7 arecollectively regarded as the second gate electrode control circuit 2801of FIG. 28.

In the second modification example, the loss of the low-side MOSFET 4006is reduced according to the condition of the load at the time of lossreduction and therefore power consumption by the semiconductorintegrated circuit device 4002 and the power supply system is reduced.

Third Modification Example

FIG. 29 is a block diagram showing a configuration of the semiconductorintegrated circuit device 4002 according to a modification example ofthe ninth embodiment. The configuration of FIG. 29 is similar to theconfiguration of FIG. 27. Differences between both configurations,therefore, will mainly be described.

In FIG. 29, 2902 denotes a single-gate MOSFET having no second gateelectrode G2, 2900 denotes a load current detecting circuit, and 2901denotes a second gate electrode control circuit.

The load current detecting circuit 2900 is identical in configurationwith the load current detecting circuit 2800 of the second modificationexample. The load current detecting circuit 2900 detects the outputcurrent (load current) Iout flowing through the output terminal T1 andsupplies a detection signal indicating whether the value of the outputcurrent Iout exceeds a given current value, to the second gate electrodecontrol circuit 2901. The second gate electrode control circuit 2901generates the second gate control signal UH according to the detectionsignal and supplies the second gate control signal UH to the second gateelectrode G2 of the high-side MOSFET 4005 via the control terminal T15.In the semiconductor integrated circuit device 4002 of FIG. 29, becausethe low-side MOSFET is the single-gate electrode MOSFET 2902, the secondgate control signal UL for the low-side MOSFET is not generated.

The configuration described in the seventh or ninth embodiment isapplied to the load current detecting circuit 2900 and the second gateelectrode control circuit 2901.

When the configuration of the seventh embodiment is applied, the loadcurrent detecting circuit 2202 of FIG. 22(A) is regarded as the loadcurrent detecting circuit 2900 of FIG. 29. The high-side voltage controlcircuit 2203, stationary resistance element 2200, and variableresistance element 2201 of FIG. 22(A) are collectively regarded as thesecond gate electrode control circuit 2901.

When the configuration of the ninth embodiment is applied, the voltageV1 is supplied to the second gate electrode G2 of the high-side MOSFET4005 in the light load case, and the voltage V2 is supplied to thesecond gate electrode G2 of the high-side MOSFET 4005 in the heavy loadcase.

In the third modification example, the loss of the high-side MOSFET 4005is reduced according to the condition of the load at the time of lossreduction and therefore power consumption by the semiconductorintegrated circuit device 4002 and the power supply system is reduced.

Tenth Embodiment

FIG. 30 is a block diagram showing a configuration of the semiconductorintegrated circuit device 4002 according to a tenth embodiment. Theconfiguration of FIG. 30 is similar to the configuration of thesemiconductor integrated circuit device 4002 of FIG. 40. In FIGS. 30 and40, the same components are denoted by the same reference numerals, anddifferent components therefore will mainly be described.

The semiconductor integrated circuit device 4002 of the tenth embodimenthas a terminal T16, which is connected to the second gate electrode G2of the low-side MOSFET 4006. As described above, the semiconductorintegrated circuit device 4002 is a package in which multiplesemiconductor chips are sealed. The terminal T16, therefore, serves asan external terminal attached to the package. A resistance element 3000is connected between the terminal (external terminal) T16 attached tothe package and the ground voltage PGND node, in which case theresistance element 3000 is outside the package.

FIG. 31 is a circuit diagram drawn by paying attention to the high-sideMOSFET 4005 and low-side MOSFET 4006 of the semiconductor integratedcircuit device 4002 of FIG. 30. Because FIG. 31 is drawn by payingattention to the high-side MOSFET 4005 and low-side MOSFET 4006, theconfiguration of the driver 4003 is omitted from FIG. 31. FIG. 31depicts parasitic resistances, parasitic capacitances, and parasiticinductances, as equivalent circuits.

In FIG. 31, the high-side MOSFET 4005 and the low-side MOSFET 4006 areidentical in configuration with each other. The low-side MOSFET 4006(high-side MOSFET 4005) has a parasitic capacitance Cgs formed betweenthe first gate electrode G1 and the source S, a parasitic capacitanceCed formed between the second gate electrode G1 and the drain D, aparasitic capacitance Cds formed between the source S and the drain D,and a parasitic diode DD formed by connecting the back gate to thesource S. Respective first gate electrodes G1 of the high-side MOSFET4005 and the low-side MOSFET 4006 are connected to the driver 4003, andare driven by the drive signals GH and GL from the driver 4003.

The drain D of the high-side MOSFET 4005 is connected to a line L1,through which the input voltage VIN from the terminal (externalterminal) T6 is input to the high-side MOSFET 4005. The line L1 isconnected to a voltage-stabilizing capacitor element Cin and isaccompanied by a parasitic inductance LP1. The drain of the high-sideMOSFET 4005 is connected to a switching node Ns via a parasiticinductance LP3, and the switching node Ns is connected to the drain D ofthe low-side MOSFET 4006. The source S of the low-side MOSFET 4006 isconnected to the ground voltage PGND node via a parasitic inductanceLP2. The switching node Ns is connected to one end of the coil element4008 of which the other end is connected to the smoothing capacitor4009. In FIG. 31, a CPU is depicted as the load 4001.

According to the tenth embodiment, the second gate electrode G2 of thehigh-side MOSFET 4005 is connected to the ground voltage PGND node(connection of the second gate electrode G2 is not limited to this). Thesecond gate electrode G2 of the low-side MOSFET 4006 is connected to theground voltage PGND node via the resistance element 3000 located outsidethe package and connected thereto via the terminal T16 (FIG. 30).According to the tenth embodiment, the parasitic capacitance Ced formedbetween the second gate electrode G2 of the low-side MOSFET 4006 and thedrain D of the low-side MOSFET 4006 and the external resistance element3000 make up a snubber circuit. This snubber circuit suppresses voltageringing at the switching node Ns.

FIGS. 32(A) to 32(E) are waveform charts showing an operation of thecircuit shown in FIG. 31. The operation of the circuit will hereinafterbe described, referring to FIGS. 31 and 32(A) to 32(E).

In FIG. 32, the horizontal axis represents time. FIG. 32(A) depicts achange in a voltage between the first gate electrode G1 and the source(Lo-Side Vgs) of the low-side MOSFET 4006, and FIG. 32(B) depicts achange in a voltage between the first gate electrode G1 and the source(Hi-Side Vgs) of the high-side MOSFET 4005. In other words, FIGS. 32(A)and 32(B) depict changes in the voltages of the drive signals GL and GHfrom the driver 4003. FIG. 32(C) depicts a current flowing through theparasitic diode DD (body diode) of the low-side MOSFET 4006 (Body DiodeForwarding Current). FIG. 32(E) depicts a voltage at the switching nodeNs that results when the snubber circuit is created by providing theexternal resistance element 3000. FIG. 32(D) depicts a voltage at theswitching node Ns that results when the snubber circuit is not created.

When the voltage Vgs between the first gate electrode G1 and the sourceof the low-side MOSFET 4006 drops in a manner as indicated in FIG.32(A), an action of the coil element 4008 causes a current to flow fromthe ground voltage PGND node through the diode DD of the low-side MOSFET4006 (FIG. 32(C)). Then, as indicated by FIG. 32(B), when the voltageVgs between the first gate electrode G1 and the source of the high-sideMOSFET 4005 rises, a sharp voltage build-up rate (dv/dt) leads to aringing phenomenon of a voltage at the switching node Ns through theparasitic inductance (FIG. 32(D)). However, by connecting the externalresistance element 3000 to the terminal T16, the resistance element 3000is connected to the second gate electrode G2 of the low-side MOSFET4006, and therefore the parasitic capacitance Ced and the externalresistance 3000 are connected in series between the ground voltage PGNDnode and the switching node Ns, thus working as the snubber circuit thatsuppresses the ringing phenomenon at the switching node Ns. As a result,as indicated by FIG. 32(E), the ringing phenomenon at the switching nodeNs is suppressed to be less intensive than the ringing phenomenon shownin FIG. 32(D). Hence a ringing phenomenon that occurs when the high-sideMOSFET 4005 is switched on is suppressed, which enables generation ofthe output voltage Vout with less noises.

According to the tenth embodiment, by adjusting the resistance value ofthe external resistance element 3000, an extent of suppression of aringing phenomenon can be adjusted. It is therefore preferable that theresistance element 3000 be connected to the terminal T16 outside thepackage, i.e., semiconductor integrated circuit device 4002.

Eleventh Embodiment

FIG. 33 is a block diagram showing a configuration of the semiconductorintegrated circuit device 4002 according to an eleventh embodiment. Theconfiguration of FIG. 33 is similar to the configuration of thesemiconductor integrated circuit device 4002 of FIG. 40. Differencesbetween both configurations, therefore, will mainly be described.

In FIG. 33, inside the semiconductor integrated circuit device 4002, thesecond gate electrode G2 of the high-side MOSFET 4005 is connected tothe ground voltage PGND node. This configuration enables a reduction innoises that are generated when the high-side MOSFET 4005 is switched on,which will be described referring to FIGS. 34 and 35.

FIG. 34 is a circuit diagram drawn by paying attention to the high-sideMOSFET 4005 and low-side MOSFET 4006 of the semiconductor integratedcircuit device 4002 of FIG. 33. Because FIG. 34 is drawn by payingattention to the high-side MOSFET 4005 and low-side MOSFET 4006, theconfiguration of the driver 4003 is omitted from FIG. 33. The circuitshown in FIG. 34 is similar to the circuit shown in FIG. 31. FIG. 34 isdifferent from FIG. 31 in that the second gate electrode G2 of thelow-side MOSFET 4006 is connected to the ground voltage PGND node withno resistance element 3000 interposed between the second gate electrodeG2 and the ground voltage PGND node and that the second gate electrodeG2 of the high-side MOSFET 4005 is connected to the ground voltage PGNDnode inside the package. The features of the circuit of FIG. 34 otherthan these two features are described in the tenth embodiment and aretherefore omitted in further description. According to the eleventhembodiment, the second gate electrode G2 of the low-side MOSFET 4006 isalso connected to the ground voltage PGND node inside the package(connection of the second gate electrode G2 of the low-side MOSFET 4006is not limited to this).

FIGS. 35(A) to 35(E) are waveform charts showing an operation of thecircuit of FIG. 34. In FIG. 35, the horizontal axis represents time.FIGS. 35(A) to 35(C) correspond to the FIGS. 32(A) to 32(C),respectively. FIG. 35(D) depicts the waveform of a current (Hi-Side Id)flowing through the source-drain path of the high-side MOSFET 4005. FIG.35(E) depicts the waveform of a voltage (Vin Ripple Voltage) at the lineL1 through which the input voltage VIN is supplied to the drain D of thehigh-side MOSFET 4005. As described referring to FIG. 31, the parasiticinductance LP1 is connected to the line L1 and the voltage stabilizingcapacitor Cin is also connected to the line L1.

According to the eleventh embodiment, inside the semiconductorintegrated circuit device 4002, the second gate electrode G2 of thehigh-side MOSFET 4005 is connected to the ground voltage PGND node. As aresult, the parasitic capacitance Ced formed between the second gateelectrode G2 and the drain is connected between the line L1 transmittingthe input voltage VIN and the ground voltage node. This parasiticcapacitance Ced is connected in parallel to the voltage-stabilizingcapacitance element Cin via the parasitic inductance LP1.

As shown in FIG. 35(B), as a voltage between the first gate electrode G1and the source of the high-side MOSFET 4005 rises and the high-sideMOSFET 4005 shifts from its off-state to on-state, an inrush currentflows because of a potential difference between the input voltage VINand the ground voltage PGND node. This inrush current appears as a peakof the drain current Id of the high-side MOSFET 4005 (FIG. 35(D)). Inresponse to this sharp change in the drain current Id, the parasiticimpedance LP1 works to oscillate a voltage at the line L1 (FIG. 35(E)).Although the voltage stabilizing capacitance Cin functions to stabilizethe voltage at the line L1, a voltage ripple still arises near the drainD of the high-side MOSFET 4005 on the line L1. According to the eleventhembodiment, the drain D of the high-side MOSFET 4005 is connected to theground voltage PGND node in the semiconductor integrated circuit device4002 through the parasitic capacitance Ced that provide AD connection.This parasitic capacitance Ced absorbs a voltage oscillation (ripple)that arises near the drain of the high-side MOSFET 4005 when thehigh-side MOSFET 4005 is switched on, thus suppressing noise generation.

The relation between the semiconductor integrated circuit device 4002,the package, and the power supply system 4000 will be described. FIG.36(A) is a block diagram showing the relation between the semiconductorintegrated circuit device 4002, the package, and the power supply system4000. The power supply system 4000 includes the control semiconductorintegrated circuit device 4007, the semiconductor integrated circuitdevice 4002, the coil element 4008, and the smoothing capacitor element4009. According to this embodiment, the semiconductor integrated circuitdevice 4002 has three semiconductor chips, which are sealed in onepackage. In this specification of the present invention, therefore, thesemiconductor integrated circuit device 4002 is equivalent to a package(denoted as 4002P in FIG. 11) having semiconductor chips (threesemiconductor chips in this embodiment) incorporated therein.

According to this embodiment, three semiconductor chips are asemiconductor chip 4005C having the high-side MOSFET 4005, asemiconductor chip 4006C having the low-side MOSFET 4006, and asemiconductor chip 4003C having the driver 4003. In FIG. 36(A), toprevent the complication of the drawing, the drive circuits 4011 and4012 are shown as functional components making up the driver 4003. Aspecific example of the driver 4003 is shown in FIG. 40. In FIG. 36(A),to clearly indicate the function of the drive circuits 4011 and 4012 ofcomplementally switching on and off the high-side MOSFET 4005 and thelow-side MOSFET 4006, the drive circuit 4011 is depicted as a bufferwhile the drive circuit 4012 is depicted as an inverter. The voltageVCIN supplied to the driver semiconductor chip 4003C and the groundvoltage CGND are omitted from FIG. 36(A).

A configuration of the package 4002P having the semiconductor chipsincorporated therein will then be described. FIG. 36(B) is a plan viewof the configuration of the package 4002P. In FIG. 36(B), P denotes eachof multiple external terminals of a lead frame and a region 3600indicated by an encircling broken line is a region sealed with a resin,etc. Out of the multiple external terminals P, given external terminalsP serves as the terminals T1 to T6 of the semiconductor integratedcircuit device 4002 described referring to FIG. 40. In FIG. 36(B), theexternal terminals P corresponding to the terminals T1, T2, and T6 areexpressed as VSWH (T1), PGND (T2), and VIN (T6), respectively.

In FIG. 36(B), 3603 denotes a tab carrying the semiconductor chip 4005Chaving the high-side MOSFET 4005, 3604 denotes a tab carrying thesemiconductor chip 4006C having the low-side MOSFET 4006, and 3605denotes a tab carrying the semiconductor chip 4003C having the driver4003. Respective terminals (pads) of the semiconductor chips 4003C,4005C, and 4006C are connected electrically to given external terminalsP or semiconductor chips through lead wires or copper sheets. FIG.36(B), for example, depicts a pad S of the source S of the high-sideMOSFET 4005, a pad G of the first gate electrode G1 of the high-sideMOSFET 4005, a pad U of the second gate electrode G2 of the high-sideMOSFET 4005, a pad S of the source S of the low-side MOSFET 4006, and apad G of the first gate electrode G1 of the low-side MOSFET 4006.

A relatively high current flows through respective source-drain paths ofthe high-side MOSFET 4005 and low-side MOSFET 4006 (current flow throughthe source-drain paths is not limited to this). For this reason, thesource S of the high-side MOSFET 4005 and the source S of the low-sideMOSFET 4006 are connected to given parts through copper sheets 3601 and3602, respectively. For example, the source S of the low-side MOSFET4006 is connected to the multiple external terminals P (T2) suppliedwith the ground voltage PGND, through the copper sheet 3602. In thisembodiment, the low-side MOSFET is constructed to be larger than thehigh-side MOSFET 4005 so that a high current can flow through the pathbetween the ground voltage PGND node and the output terminal T1.

According to the eleventh embodiment, inside the semiconductorintegrated circuit device 4002, the second gate electrode G2 of thehigh-side MOSFET 4005 is connected to the ground voltage PGN node. InFIG. 36(B), the second gate electrode G2 of the high-side MOSFET 4005corresponds to the pad U. As shown in FIG. 36(B), therefore, in thepackage region indicated by the broken line 3600, the pad U of thehigh-side MOSFET 4005 is connected to an external terminal P suppliedwith the ground voltage PGND, through a lead wire 3606.

In the tenth embodiment, a pad to which the second gate electrode G2 ofthe low-side MOSFET 4006 is connected is connected to a given externalterminal P, and outside the package region indicated by the broken line3600, the resistance element 3000 is connected to the external terminalP.

The present invention is not limited to the above embodiments but mayinclude various modification examples. The above first to eleventhembodiments are described in detail for facilitating understanding ofthe present invention and each embodiment does not necessarily includeall the constituent elements described above. Part of the configurationof one embodiment may be replaced with the configuration of anotherembodiment. The configuration of another embodiment may be added to theconfiguration of one embodiment. Part of the configuration of eachembodiment may be deleted or replaced with the configuration of anotherembodiment or the configuration of another embodiment may be added topart of the configuration of each embodiment.

For example, the configuration of the eleventh embodiment may be addedto the configuration of the tenth embodiment. The first to fourthembodiments and the second modification example of the ninth embodimentrelate to the low-side MOSFET. The configuration of the eleventhembodiment, therefore, may be added to the configurations of the firstto fourth embodiments and the second modification example of the ninthembodiment. The fifth to eighth embodiments and the first modificationexample of the ninth embodiment relate to the high-side MOSFET. Theconfiguration of the tenth embodiment, therefore, may be added to theconfigurations of the fifth to eighth embodiments and the firstmodification example of the ninth embodiment.

In the first, second, and tenth embodiments, the high-side MOSFET may beprovided as a single-gate electrode MOSFET. In the fourth to sixth andeleventh embodiments, the low-side MOSFET may be provided as asingle-gate electrode MOSFET.

The above embodiments are described as cases where the high-side MOSFETand low-side MOSFET are provided as n-channel type MOSFETs. It isobvious, however, that both MOSFET may be provided as p-channel typeMOSFETs.

The specification of the present invention discloses multipleinventions, inventions, some of which are described as claims. Thespecification, however, also discloses other inventions, out of whichtypical inventions are described below.

(A) A semiconductor integrated circuit device that has a first voltageterminal, a second voltage terminal supplied with a voltage lower than avoltage supplied to the first voltage terminal, and an output terminaland that cyclically changes the direction of a current supplied to acoil element connected to the output terminal, the semiconductorintegrated circuit device including:

a first MOSFET having a first input electrode, a drain, and a source,the first MOSFET being connected between the first voltage terminal andthe output terminal and electrically connecting the first voltageterminal to the output terminal according to a first input signalsupplied to the first input electrode;

a second MOSFET having a first input electrode, a drain, a source, and asecond input electrode located closer to the drain than the first inputelectrode, the second MOSFET being connected between the second voltageterminal and the output terminal and electrically connecting the secondvoltage terminal to the output terminal according to a second inputsignal supplied to the first input electrode;

a drive circuit connected to respective first input electrodes of thefirst MOSFET and the second MOSFET, the drive circuit generating thefirst input signal and the second input signal so that the first MOSFETand the second MOSFET are switched on and off complementally; and

an external terminal to which a second input electrode of the secondMOSFET is connected,

in which the first MOSFET, the second MOSFET, and the drive circuit aresealed in one package, the external terminal is attached to the package,and a resistance element is connected between the external terminal anda given voltage node.

(B) The semiconductor circuit device according to the item (A), in whichthe second MOSFET has a first conductive first semiconductor region, asecond conductive second semiconductor region overlaid on the firstsemiconductor region, and a first conductive third semiconductor regionoverlaid on the second semiconductor region; and

a drain of the second MOSFET is composed of the first semiconductorregion, a source of the second MOSFET is composed of the thirdsemiconductor region, a first input electrode of the second MOSFET iscomposed of a first metal layer embedded in the second semiconductorregion across an insulating layer, and a second input electrode of thesecond MOSFET is composed of a second metal layer embedded in the firstsemiconductor region across an insulating layer.

(C) A semiconductor integrated circuit device that has a first voltageterminal, a second voltage terminal supplied with a voltage lower than avoltage supplied to the first voltage terminal, and an output terminaland that cyclically changes the direction of a current supplied to acoil element connected to the output terminal, the semiconductorintegrated circuit device including:

a first MOSFET having a first input electrode, a drain, a source, and asecond input electrode located closer to the drain than the first inputelectrode, the first MOSFET being connected between the first voltageterminal and the output terminal and electrically connecting the firstvoltage terminal to the output terminal according to a first inputsignal supplied to the first input electrode;

a second MOSFET having a first input electrode, a drain, and a source,the second MOSFET being connected between the second voltage terminaland the output terminal and electrically connecting the second voltageterminal to the output terminal according to a second input signalsupplied to the second input electrode; and

a drive circuit connected to respective first input electrodes of thefirst MOSFET and the second MOSFET, the drive circuit generating thefirst input signal and the second input signal so that the first MOSFETand the second MOSFET are switched on and off complementally,

in which the first MOSFET, the second MOSFET, and the drive circuit aresealed in one package, and the second input electrode of the firstMOSFET is connected to the second voltage terminal in the package.

(D) The semiconductor integrated circuit device according to the item(C),

in which the first MOSFET has a first conductive first semiconductorregion, a second conductive second semiconductor region overlaid on thefirst semiconductor region, and a first conductive third semiconductorregion overlaid on the second semiconductor region, and

wherein a drain of the first MOSFET is composed of the firstsemiconductor region, a source of the first MOSFET is composed of thethird semiconductor region, a first input electrode of the first MOSFETis composed of a first metal layer embedded in the second semiconductorregion across an insulating layer, and a second input electrode of thesecond MOSFET is composed of a second metal layer embedded in the firstsemiconductor region across an insulating layer.

(E) A semiconductor integrated circuit device including:

a first voltage terminal supplied with a first voltage;

a second voltage terminal supplied with a second voltage different involtage value from the first voltage;

an output terminal;

a first MOSFET having a first input electrode, a drain, and a source,the first MOSFET being connected between the first voltage terminal andthe output terminal and electrically connecting the first voltageterminal to the output terminal according to a first input signalsupplied to the first input electrode;

a second MOSFET having a first input electrode, a drain, a source, and asecond input electrode located closer to the drain than the first inputelectrode, the second MOSFET being connected between the second voltageterminal and the output terminal and electrically connecting the secondvoltage terminal to the output terminal according to a second inputsignal supplied to the first input electrode;

a drive circuit connected to respective first input electrodes of thefirst MOSFET and the second MOSFET, the drive circuit generating thefirst input signal and the second input signal so that the first MOSFETand the second MOSFET are switched on and off complementally;

a first voltage generating circuit connected to the second inputelectrode of the second MOSFET, the first voltage generating circuitsupplying a negative voltage negative in polarity relative to a voltageat the source of the second MOSFET, to the second input electrode;

a detecting circuit that detects a current flowing through the outputterminal; and

a control circuit that supplies a control signal following a detectionsignal from the detecting circuit, to the second input electrode of thesecond MOSFET.

(F) A semiconductor integrated circuit device including:

a first voltage terminal supplied with a first voltage;

a second voltage terminal supplied with a second voltage different involtage value from the first voltage;

an output terminal;

a first MOSFET having a first input electrode, a drain, and a source,the first MOSFET being connected between the first voltage terminal andthe output terminal and electrically connecting the first voltageterminal to the output terminal according to a first input signalsupplied to the first input electrode;

a second MOSFET having a first input electrode, a drain, a source, and asecond input electrode located closer to the drain than the first inputelectrode, the second MOSFET being connected between the second voltageterminal and the output terminal and electrically connecting the secondvoltage terminal to the output terminal according to a second inputsignal supplied to the first input electrode;

a drive circuit connected to respective first input electrodes of thefirst MOSFET and the second MOSFET, the drive circuit generating thefirst input signal and the second input signal so that the first MOSFETand the second MOSFET are switched on and off complementally; and

a control circuit that changes a voltage supplied to the second gateelectrode of the second MOSFET, in synchronization with switching on/offof the second MOSFET.

EXPLANATION OF REFERENCE NUMERALS

-   -   1000, 1600, 2701, 2801, 2901 Second gate electrode control        circuit    -   4000 Power supply system    -   4002 Semiconductor integrated circuit device    -   4003 Driver    -   4004 Control circuit    -   4005 High-side MOSFET    -   4006 Low-side MOSFET    -   4007 Control semiconductor integrated circuit device    -   G1 First gate electrode    -   G2 Second gate electrode    -   GL, GH Drive signal    -   UH, UL Second gate control signal

1-20. (canceled)
 21. A semiconductor integrated circuit devicecomprising: a first voltage terminal supplied with a first voltage; afirst MOSFET having: a drain; a source; a first input electrodeconfigured to cause a current path to be formed between the source andthe drain; and a second input electrode located closer to the drain thanthe first input electrode and aligned beside the current path, the firstinput electrode partially overlapping with the drain and insulated fromthe drain and the second input electrode by an insulating material, thedrain being electrically connected to the first voltage terminal; afirst resistance element electrically connected to the source and aconnection node; and a second resistance element electrically connectedto the connection node and a ground terminal, which is supplied with asecond voltage lower than the first voltage, wherein the connection nodeis electrically connected to the second input electrode.
 22. Thesemiconductor integrated circuit device according to claim 21, whereinthe first MOSFET has a first-conductivity-type first semiconductorregion, a second-conductivity-type second semiconductor region overlaidon the first semiconductor region, and a first-conductivity-type thirdsemiconductor region overlaid on the second semiconductor region, thedrain of the first MOSFET is composed of the first semiconductor region,the source of the first MOSFET is composed of the third semiconductorregion, the first input electrode of the first MOSFET is composed of afirst-conductivity-type material embedded in the second semiconductorregion, and the second input electrode of the first MOSFET is composedof a second-conductivity-type material embedded in the firstsemiconductor region.
 23. The semiconductor integrated circuit deviceaccording to claim 21, wherein the source is electrically connected toan output terminal for connecting to an external load.
 24. Thesemiconductor integrated circuit device according to claim 21, whereinthe first MOSFET, the first resistance element, the second resistanceelement and the connection node are sealed in one package.
 25. Thesemiconductor integrated circuit device according to claim 21, furthercomprising a drive circuit that outputs a drive signal to the firstinput electrode.
 26. The semiconductor integrated circuit deviceaccording to claim 25, wherein the first MOSFET, the first resistanceelement, the second resistance element, the connection node, and thedrive circuit are sealed in one package.
 27. The semiconductorintegrated circuit device according to claim 26, wherein the drivecircuit is formed in a first semiconductor chip, the first MOSFET isformed in a second semiconductor chip different from the firstsemiconductor chip, and the first and second semiconductor chips aresealed in the one package.
 28. A method of driving a semiconductorintegrated circuit device comprising: providing a semiconductorintegrated circuit device including: a first voltage terminal; a firstMOSFET having: a drain electrically connected to the first voltageterminal; a source; a first input electrode configured to cause acurrent path to be formed between the source and the drain; and a secondinput electrode located closer to the drain than the first inputelectrode and aligned beside the current path, the first input electrodepartially overlapping with the drain and insulated from the drain andthe second input electrode by an insulating material; a first resistanceelement electrically connected to the source; a second resistanceelement electrically connected to a ground terminal; and a connectionnode electrically connected to the first resistance element, the secondresistance element, and the second input electrode; supplying a firstvoltage to the first voltage terminal; supplying a second voltage to theground terminal; and inputting a first input voltage to the first inputelectrode, wherein an amplitude of a second input voltage supplied tothe second input electrode is smaller than an amplitude of voltage ofthe source, and the second input voltage follows the voltage of thesource.
 29. The method of driving a semiconductor integrated circuitdevice according to claim 28, wherein the second input voltage isnegative in polarity relative to the voltage of the source.
 30. Themethod of driving a semiconductor integrated circuit device according toclaim 28, wherein, when the first MOSFET in its off-state is switchedon, a voltage rising in a same manner as the voltage of source issupplied to the second input electrode.
 31. The method of driving asemiconductor integrated circuit device according to claim 28, wherein,when the first MOSFET in its on-state is switched off, the second inputvoltage is lower than the voltage of source and is negative in polarityrelative to the voltage of source.